Magnetic element in a non-saturated region in a trnsformer/ inductor

ABSTRACT

Single, multistage, and distributed magnetic switched and tank resonant power conversion systems can use a transformer with a magnetic element operating in a non-saturated region (NSME). The NSME provides superior protection to conducted lightning transients, superior thermal operating bandwidth, higher magnetizing efficiency, greater flux/power density potential and form factor flexibility when implemented with many circuit strategies, some of which are disclosed herein. Efficient energy storage and transfer is achieved by the optimized application of NSME. The use of efficient rectifying flyback management techniques protects switches and provides additional output. The preferred embodiment teaches against common theories of transformers and uses a NSME in the core.

CROSS REFERENCE APPLICATIONS

[0001] This application is a divisional application of application Ser.No. 09/923,027 filed 08/06/2001 which is a divisional of applicationSer. No. 09/410,849 filed 10/01/1999 and issued as U.S. Pat. No6,272,025 on 08/07/2001.

FIELD OF INVENTION

[0002] The present invention relates to converters, power supplies, moreparticularly, to single, or multi stage, AC/DC or DC/DC isolated andnon-isolated push-pull converters including but not limited to, forward,flyback, buck, boost, push pull, and resonant mode converters, and powersupplies, having individual or distributed NSME with high speed FETswitching and efficient flyback management and or having input PFC(power factor correction) and input protection from lightningtransients. The invention also allows the magnetic element(s) bedistributed to accommodate packaging restrictions, multiple secondarywindings, or operation at very high winding voltages.

BACKGROUND OF THE INVENTION

[0003] There are several basic topologies commonly used to implementswitching converters.

[0004] A DC-DC converter is a device that converts a DC voltage at onelevel to a DC voltage at another level. The converter typically includesa magnetic element having primary and secondary windings wound around itto form a transformer. By opening and closing the primary circuit atappropriate intervals control over the energy transfer between thewindings occurs. The magnetic element provides an alternating voltageand current whose amplitude can be adjusted by changing the number andratio of turns in each set of the windings. The magnetic elementprovides galvanic isolation between the input and the output of theconverter.

[0005] One of the topologies is the push-pull converter. The outputsignal is the output of an IC network that switches the transistorsalternately “on” and “off”. High frequency square waves on thetransistor output drive the magnetic element into AC (alternatingcurrent) bias. The isolated secondary outputs a wave that is rectifiedto produce DC (direct current). The push-pull converters generally havemore components as compared to other topologies. The push-pull approachmakes efficient use of the magnetic element by producing AC bias, butsuffers from high parts count, thermal derating, oversized magnetics,and elaborate core reset schemes. The destructive fly-back voltagesoccurring across the switches are controlled through the use ofdissipative snubber networks positioned across the primary switches.Another of the topologies is the forward converter. When the primary ofthe forward converter is energized, energy is immediately transferred tothe secondary winding. In addition to the aforementioned issues theforward converter suffers from inefficient (dc bias) use of the magneticelement. The prior art power supplies use high permeability gappedferrite magnetic elements. These are well known in the art and arewidely used. The magnetics of the prior art power supplies are generallydesigned for twice the required power rating and require complex methodsto reset and cool the magnetic elements resulting in increased costs andlimited operating temperatures. This is because high permeabilitymagnetic elements saturate during operation producing heat in the core,which increases permeability and lowers the saturation threshold. Thisproduces runaway heating, current spikes and/or large leakage currentsin the air gap, reduced efficiency, and ultimately less power at highertemperatures and/or high load. The overall effects are, lowerefficiency, lower power density, and forced air/heatsink dependantsupplies that require over-rated ferrite magnetic elements for a givenoutput over time, temperature, and loading.

Improvements

[0006] The combined improvements of the invention translate to highersystem efficiencies, higher power densities, lower operatingtemperatures, and, improved thermal tolerance thereby reducing oreliminating the need for forced air cooling per unit output. Thenon-saturating magnetic properties are relatively insensitive totemperature (see FIG. 17), thus allowing the converter to operate over agreater temperature range. In practice, the operating temperature forthe NSME is limited to 200 C. by wire/core insulation; thenon-saturating magnetic material remains operable to near its Curietemperature of 500 C. What are needed are converters having circuitstrategies that make advantageous use of individual and distributedNSME.

[0007] What are needed are converters having buffer circuits thatprovide fast, low impedance critically damped switching of the mainFET's.

[0008] What are needed are converters that incorporate efficientmultiple “stress-less” flyback management techniques to rectify andcritically damp excessive node voltages across converter switches.

[0009] What are needed are converters having flux feedback frequencymodulation.

[0010] What are needed are converters that correct AC power factor.

[0011] What is needed are converters that meet or exceed class Bconducted EMI requirements.

[0012] What are needed are converters tolerant of lightning and harshthermal environments. The present invention addresses these and more.

SUMMARY OF THE INVENTION

[0013] The main aspect of the present invention is to implementconverters having circuit strategies that make advantageous use ofindividual and distributed NSME for the achievement of the keyperformance enhancements disclosed herein.

[0014] Another aspect of the present invention is to provide uniqueresonant tank circuit converter strategies with individual anddistributed NSME that make use of higher primary circuit voltageexcursions in the production of high frequency/high density magneticflux.

[0015] Another aspect of the present invention is a high energy densitysingle stage frequency controlled resonant tank converter topologyenabled by the use of individual and distributed NSME. Another aspect ofthe present invention is to provide a converter design that utilizes aFET drive technique consisting of an ultra fast, low RDS on N-channelFET for charging the main FET gate and an ultra fast P-channeltransistor for discharging the main FET gate.

[0016] Another aspect of the present invention is to provide convertersthat incorporate efficient multiple “stressless” flyback managementtechniques to rectify and critically damp excessive node voltages acrossconverter switches.

[0017] Another aspect of the present invention is to provide a converterhaving core (flux) synchronized zero crossing frequency modulation.

[0018] Another aspect of the present invention is to present a highpower factor to the AC line.

[0019] Another aspect of the present invention is to provide

[0020] Another aspect of the present invention is to provide protectionfrom high voltage (input line) transients.

[0021] Another aspect of the present invention is to combine distributedmagnetics advantageously with the other converter aspects.

[0022] Another aspect of the present invention is active ripplerejection provided by several high-gain high-speed isolated control andfeedback systems.

[0023] Other aspects of this invention will appear from the followingdescription and appended claims, reference being made to theaccompanying drawings forming a part of this specification wherein likereference characters designate corresponding parts in the several views.

BRIEF DESCRIPTION OF THE DRAWINGS

[0024]FIG. 1 and 1A is a schematic diagram of a two-stage power factorcorrected AC to DC isolated output converter embodiment of theinvention.

[0025]FIG. 2 is a schematic diagram of a single stage DC to AC converterembodiment with isolated output sub-circuit DCAC1.

[0026]FIG. 3 and 3A is a schematic diagram of a three stage AC to DCisolated output converter embodiment of the invention.

[0027]FIG. 4 is a schematic diagram of a power factor corrected singlestage AC to DC converter sub-circuit ACDFPF.

[0028]FIG. 5 is a graph comparing typical winding currents ininductance.

[0029]FIG. 6 is a schematic for a non-isolated low side switch buckconverter sub-circuit NILBK.

[0030]FIG. 7 is the preferred embodiment schematic for a tank coupledsingle stage converter sub-circuit TCSSC.

[0031]FIG. 8 is a schematic for a tank coupled totem pole convertersub-circuit TCTP.

[0032]FIG. 9 is a block diagram for a single stage non-isolated DC to DCboost converter NILSBST.

[0033]FIG. 10 is a schematic for a two stage isolated DC to DC boostcontrolled push-pull converter BSTPP.

[0034]FIG. 11 is a graph of permeability as a function of temperaturefor typical prior art magnetic element material.

[0035]FIG. 12 is a graph of flux density as a function of temperaturefor typical prior art magnetic element material.

[0036]FIG. 12A is a graph of magnetic element losses for various fluxdensities and operating frequencies typical of prior art magneticelement material.

[0037]FIG. 13 is a graph showing standard switching losses.

[0038]FIG. 14 is a graph showing lower switching losses of theinvention.

[0039]FIG. 15 is a graph showing the magnetizing curve (BH) for the NSMEmaterial.

[0040]FIG. 16 is a graph of magnetic element losses for various fluxdensities and operating frequencies of the NSME material.

[0041]FIG. 17 is a graph of permeability as a function of temperaturefor the NSME.

[0042]FIG. 18 is a schematic representation of the boost NSMEsub-circuit PFT1.

[0043]FIG. 18A is a schematic representation of the NSME sub-circuitPFT1A.

[0044]FIG. 18B is a schematic representation of the non-saturating twoterminal NSME sub-circuit BL1.

[0045]FIG. 18C is a schematic diagram of the NSME implemented asdistributed magnetic assembly PFT1D.

[0046]FIG. 19 is a schematic representation of the push-pull NSMEsub-circuit PPT1.

[0047]FIG. 19A is a schematic representation of the alternate push-pullNSME sub-circuit PPT1A.

[0048]FIG. 20 is a schematic diagram of the NSME input transientprotection and line filter sub-circuit LL.

[0049]FIG. 21 is a schematic diagram of the alternate NSME inputtransient protection and line filter sub-circuit LLA.

[0050]FIG. 22 is a schematic diagram of the AC line rectifiersub-circuit BR.

[0051]FIG. 23 is a schematic diagram of the power factor controllersub-circuit PFA.

[0052]FIG. 24 is a schematic diagram of the alternate power factorcorrecting boost control element sub-circuit PFB.

[0053]FIG. 25 is a schematic diagram of the output rectifier and filtersub-circuit OUTA.

[0054]FIG. 25A is a schematic diagram of an alternate rectifiersub-circuit OUTB.

[0055]FIG. 25B is a schematic diagram of an alternate final outputrectifier and filter sub-circuit OUTBB.

[0056]FIG. 26 is a schematic diagram of the floating 18_Volt DC controlpower sub-circuit CP.

[0057]FIG. 27 is a schematic diagram of the alternate floating 18_VoltDC push-pull control power sub-circuit CPA.

[0058]FIG. 28 is a schematic diagram of the over temperature protectionsub-circuit OTP.

[0059]FIG. 29 is a schematic diagram of the high-speed low impedancebuffer sub-circuit AMP, AMP1, AMP2 and AMP3.

[0060]FIG. 30 is a schematic diagram of the main switch snubbersub-circuit SN.

[0061]FIG. 30A is a schematic diagram of the main switch rectifyingdiode snubber sub-circuit DSN.

[0062]FIG. 31 is a schematic diagram of the alternate snubbersub-circuit SNA.

[0063]FIG. 32 is a schematic diagram of the mirror snubber sub-circuitSNB.

[0064]FIG. 33 is a schematic diagram of the pulse-width/Frequencymodulator sub-circuit PWFM.

[0065]FIG. 34 is an oscillograph of node voltages measured duringoperation of sub-circuit PWFM (FIG. 33).

[0066]FIG. 35 is an oscillograph of the primary tank voltage measuredduring operation of sub-circuit TCTP (FIG. 8).

[0067]FIG. 36 is a schematic diagram of the non-isolated 18-Volt DCcontrol power sub-circuit REG.

[0068]FIG. 37 is a schematic for a non-isolated high-side switch buckconverter sub-circuit HSBK.

[0069]FIG. 38 is a schematic for the low-side buck regulated two-stageconverter embodiment with isolated push-pull output sub-circuit LSBKPP.

[0070]FIG. 39 is a schematic for an alternate isolated two-stagelow-side switch buck converter sub-circuit LSBKPPBR.

[0071]FIG. 40 is a schematic diagram of the over voltage feed backsub-circuit IPFFB.

[0072]FIG. 40A is a schematic diagram of the non-isolated boost outputvoltage feedback sub-circuit FBA.

[0073]FIG. 40B is a schematic diagram of the isolated output voltagefeedback sub-circuit IFB.

[0074]FIG. 40C is a schematic diagram of the alternate isolated overvoltage feedback sub-circuit IOVFB.

[0075]FIG. 41 is a schematic diagram of the non-isolated output voltagefeedback sub-circuit FBI.

[0076]FIG. 42 is a schematic diagram of an over voltage protectionsub-circuit OVP.

[0077]FIG. 42A is a schematic diagram of the isolated over voltagefeedback sub-circuit OVP1.

[0078]FIG. 42B is a schematic diagram of the over voltage protectionsub-circuit OVP2.

[0079]FIG. 42C is a schematic diagram of the isolated over voltagefeedback sub-circuit OVP3.

[0080]FIG. 43 is a schematic diagram of the Push-pull oscillatorsub-circuit PPG.

Before explaining the disclosed embodiments of the present invention indetail, it is to be understood that:

[0081] The invention is not limited in its application to the details ofthe particular arrangements shown or described, since the invention iscapable of other embodiments. since the invention is capable of otherembodiments.

[0082] The expression “distributed magnetic(s)” refers to theconfiguration of multiple magnetic elements that share a single seriescoupled primary winding to induce isolated output currents from multipleseries or parallel secondary windings.

[0083] Also, the terminology used herein is for the purpose ofdescription not limitation.

DESCRIPTION OF THE PREFERRED EMBODIMENT

[0084] In this and other descriptions contained herein, the followingsymbols shall have the meanings attributed to them: “+” shall indicate aseries connection, such as resistor A in series with resistor B shown as“A+B”. “∥” Shall indicate a parallel connection, such as resistor A inparallel with resistor B shown as “A∥B”.

[0085] Referring first to FIG. 7, a schematic diagram of the preferredembodiment of the invention.

[0086]FIG. 7 is a schematic of the preferred embodiment of a tankcoupled single stage converter sub-circuit TCSSC. Sub-circuit TCSSCconsists of resistor R20 and RLOAD, capacitor C10, transistors Q21 andQ11, sub-circuit CP (FIG. 26), sub-circuit PFT1 (FIG. 18), sub-circuitOUTA (FIG. 25), sub-circuit AMP (FIG. 29), sub-circuit IFB (FIG. 40B)and sub-circuit PWFM (FIG. 33). Table number R20 1k ohms R61 2k ohms Q21TST541 U12 4N29 Q11 IRFP460 C10 1.8 uf

[0087] TCSSC can be configured to operate as an AC-DC converter, a DC-DCconverter, a DC-AC converter, and an AC-AC converter. Sub-circuit TCSSCconsists of resistor R20 and RLOAD, capacitor C10, switches Q11 and Q21,opto-isolator U12, sub-circuit PFT1 (FIG. 18), sub-circuit OUTA (FIG.25), sub-circuit CP (FIG. 26), sub-circuit AMP (FIG. 29), sub-circuitIFB (FIG. 40B) and sub-circuit PWFM (FIG. 33). External power sourceVBAT connects to pins DCIN+ and DCIN−. Source power may also be derivedfrom rectified AC line voltage such as FIG. 20 or FIG. 21 to form asingle stage power factor corrected AC to DC converter with isolatedoutput. From DCIN+ resistor R20 connects to sub-circuit CP pin CP+,sub-circuit AMP pin GA+, U12 LED anode and to sub-circuit PWFM pinPWFM+. Resistor R20 provides startup power to the converter until thecontrol supply regulator sub-circuit CP reaches the desired 18-voltoutput. VBAT negative is the ground return node connects to sub-circuitPWFM pin PWFM0, Q11 source, sub-circuit AMP pin GA0, sub-circuit CP pinCT0, pin DCIN− and sub-circuit PFT1 pin S1CT. Magnetic element windingnode S1H of sub-circuit PFT1 is connected to CP pin CT1A. Magneticelement winding node S1L of sub-circuit PFT1 is connected to CP pinCT2A. Sub-circuit PWFM is designed as a constant 50% duty-cycle variablefrequency generator. Sub-circuit PWFM Clock output pin CLK is connectedto input of buffer sub-circuit AMP pin GA1. The output of buffersub-circuit AMP pin GA2 is connected to the gate of Q11 and R21.Resistor R21 is connected to the cathode of U12 LED. The emitter of Q21and drain of Q11 is connected to sub-circuit PFT1 pin P1A. Pin P1B ofsub-circuit PFT1 is connected through tank capacitor C10 to node DCIN+,Q21 collector and through resistor R61 to U12 phototransistor collector.The emitter of U12 phototransistor is connected to the base of Q21. WithPWFM pin CLK high transistor Q11 conducts charging capacitor C10 throughNSME PFT1 from VBAT storing energy in PFT1. Sub-circuit PWFM switchesCLK low, Q11 turns “off”. With CLK low LED of U12 is turned “on”injecting base current into Q21. With transistor Q21 “on” the tankcircuit is completed, allowing capacitor C10 to discharge into NSME PFT1winding 100 (FIG. 18). Now the energy not transferred into the load isreleased from NSME PFT1 into the now forward biased NPN switch Q21 backinto capacitor C10. Thus any energy not used by the secondary loadremains in the tank coupled primary circuit (winding 100). When theswitching occurs at the resonant frequency, high voltages oscillatebetween C10 and winding 100 creating high flux density AC excursions inPFT1. C10 and PFT1 exchange variable AC currents whose magnitude iscontrolled by frequency modulation scheme IFB and PWFM. The largeprimary voltage generates large, high frequency biases in the NSME PFT1thereby producing high flux density AC excursions to be harvested bysecondary windings 102 and 103 (FIG. 18) to support a load or rectifiersub-circuit OUTA. Magnetic element winding node S2H of sub-circuit PFT1is connected to OUTA pin C7B. Magnetic element winding node S2L ofsub-circuit PFT1 is connected to OUTA C8B. Magnetic element winding nodeS2CT of sub-circuit PFT1 is connected to OUTA pin OUT−. Node OUT− isconnected to RLOAD, pin B− and to sub-circuit IFB pin OUT−. Rectifiedpower is delivered to pin OUT+ of OUTA and is connected to RLOAD, pin B+and to sub-circuit IFB pin OUT+. Sub-circuit IFB provides the isolatedfeedback signal to the sub-circuit PWFM. Frequency control pin FM1 ofsub-circuit PWFM is connected to sub-circuit IFB pin FBE. Internalreference pin REF of sub-circuit PWFM is connected to sub-circuit IFBpin FBC. PWFM is designed to operate at the resonate frequency of thetank (2*pi*(square root (Co*inductance of 100 (FIG. 18)). Whensub-circuit IFB senses the converter output is at the target voltage,current from PWFM pin REF is injected into FM1. Injecting current intoFM1 commands the PWFM to a lower clock frequency pin CLK. Driving thetank out of resonance reduces the amount of energy added to the tankthus reducing the converter output voltage. In the event the feedbacksignal from IFB commands the PWFM off or 0 Hz, i.e.: at no load, allprimary activity stops. The input current from VBAT may be steady stateor variable DC. When TCSSC is operated from rectified AC (sub-circuit LLFIG. 20), high input (line) power factor and input transient protectionis achieved. The primary and secondary currents of PFT1 are sinusoidaland free of edge currents of PFT1 are sinusoidal and free of edgetransitions making the converter very quiet. In addition the switchesQ11 and Q21 are never exposed to the large circulating voltage inducedin the tank (See FIG. 35). This allows the use of lower voltage switchesin the design thereby reducing losses and increasing the MTBF.Sub-circuit TCSSC takes advantage of the desirable properties of theNSME in this converter topology. TCSSC is well suited for implementationwith distributed NSME PFT1D (FIG. 18C). This combination exemplifies howdistributed magnetics enable advantageous high voltage converter designvariations that support form factor flexibility and multiple parallelsecondary outputs from series coupled voltage divided primary windingsacross multiple NSME. This magnetic strategy is useful in addressingwire/core insulation, form factor and packaging limitations, circuitcomplexity and manufacturability. These converter strategies are veryuseful for obtaining isolated high current density output from a highvoltage low current series coupled primary. Adjusting the secondaryturn's ratio allows TCSSC to generate very large AC or DC outputvoltages as well as low-voltage high current outputs.

Additional Embodiments

[0088]FIG. 1 and 1A is a schematic diagram of a two stage power factorcorrected AC to DC converter. The invention is comprised of lineprotection filter sub-circuit LL (FIG. power factor corrected regulatedboost stage with sub-circuits PFA2 (FIG. 23), snubber sub-circuit SN(FIG. 30), magnetic element sub-circuit PFT1 (FIG. 18), sub-circuit CP(FIG. 26), buffer sub-circuit AMP (FIG. 29), over temperaturesub-circuit OTP (FIG. 28), over voltage feedback sub-circuit IPFFB (FIG.40) and voltage feedback sub-circuit IFB (FIG. 40B). Start up resistorR2, filter capacitor C1, PFC capacitor C2, flyback diode D4, switchtransistor Q1, hold up capacitors C17 and C16, and resistor R17. Anefficient push-pull isolation stage with sub-circuits CPA (FIG. 27), PPG(FIG. 43), AMP1 (FIG. 29), AMP2 (FIG. 29), snubber sub-circuits SNB(FIG. 32) and SNA (FIG. 31), resistor Rload, transistors Q6 and Q9,magnetic element PPT1 (FIG. 19), and OUTA (FIG. 25). Table ElementValue/part number Cl 0.01 uf C2 1.8 uf R2 100k ohms D4 8A,600 V Q1 IRFP460 C17 100 uf C16 100 uf R17 375k ohms Q6 FS 14SM-18 A Q9 FS 14SM-18 A

[0089] In the two-stage converter the primary side voltage to the secondpush-pull output stage is modulated by the power factor corrected input(boost) stage. Each stage can comprise of individual and distributedNSME. A graph of B-H hysteresis for the non-saturating magnetics is setforth in FIG. 15. Although the following description is in terms ofparticular converter topologies, i.e., flyback controlled primary andconstant duty cycle push pull secondary, number of outputs, the style,and arrangement of the several topologies are offered by way of example,not limitation. In addition non-saturating magnetics BL1, PFT1, and PPT1may be implemented as distributed NSME. As an example PFT1 is shown as adistributed magnetic PFT1A (FIG. 18C). Distributed magnetics enableadvantageous high voltage converter design variations that support formfactor flexibility and multiple parallel secondary outputs from seriescoupled voltage divided primary windings across multiple NSME. Thenegative of the hold-up capacitor(s) [C171∥C16] is connected to bridgepositive. This allows the rectified line voltage to be excluded from theboost voltage in the hold-up capacitor(s). This, in turn, allows directregulation of the push-pull stage from the boost (PFC) stage. Thiseliminates the typical PWM control of the oversized thermally deratedtransformer and many sub-circuit components from the known art. AC lineis connected to sub-circuit LL (FIG. 20) between pins LL1 and LL2.AC/earth ground is connected to node LL0. The filtered and voltagelimited AC line appears on node/pin LL5 of sub-circuit LL and connectedto node BR1 of bridge rectifier sub-circuit BR (FIG. 22). The neutral/ACreturn leg of the filtered and voltage limited AC appears on pin LL6 ofsub-circuit LL is connected to input pin BR2 of BR. The line voltage isfull-wave rectified and is converted to a positive haversine appearingon node BR+ of sub-circuit BR (FIG. 22). Start up resistor R2 connectsBR+ to sub-circuit CP pin CP+. Node CP+ connects to pins PFA+ of controlelement sub-circuit PFA (FIG. 23) and over temperature switchsub-circuit OTP (FIG. 28) pin GAP. Resistor R2 provides start up powerto the control element until the rectifier/regulator CP is at fulloutput. Node S1H from PFT1 is connected to node PFVC of sub-circuit PFA.The zero crossings of the core are sensed when the voltage at S1H is atzero. The core zero crossings are used to reset the PFC and start a newcycle. The positive node of the DC side of bridge BR+ is connectedthrough capacitor C2 to BR−. C2 is selected for various line and loadconditions to de-couple switching current from the line improving powerfactor while reducing line harmonics and EMI. Primary of NSMEsub-circuit PFT1 (FIG. 18) pins P1B and S2CT connects to pin SNL1 ofsnubber sub-circuit SN (FIG. 30), to sub-circuit BR pin BR+ and connectsto pin BR+ (FIG. 1A). The return line for the rectified AC power BR- isconnected to the following pins: BR− of sub-circuit BR, PFA pin BR−,sub-circuit AMP pin GA0, output switch Q1 source, capacitor C2,sub-circuit CP pin CT0 sub-circuit PFT1 pin SICT and CT20 through EMIfilter capacitor C1 to earth ground node LL0. Pin BR+ from FIG. 1 isconnected to FIG. 1A sub-circuits CPA pin SN pin SNL1, sub-circuit PFT1pin P1B, and sub-circuit PFT1 pin S2CT. Pin BR+ continues to FIG. 1Aconnecting to sub-circuit CPA pin CT20, PPG (FIG. 43) pin PPG0,sub-circuit AMP1 pin GA0, sub-circuit AMP2 pin GA0, sub-circuit IPFFBpin PF−, Capacitor [C16∥C17∥resistor R17], transistor Q6 source,transistor Q9 source, sub-circuit SNA pin SNA2 and sub-circuit SNB pinSNB2. The drain of output switch Q1 is connected to diode D4 anode,sub-circuit SNB pin SNL2, and sub-circuit PFT1 pin P1A and sub-circuitSN pin SNL2. Snubber network SN reduces the high voltage stress to Q1until flyback diode D4 begins conduction. Line coupled, power factorcorrected boost regulated output voltage of the AC to DC converter stage(FIG. 1) appears on node PF+. Addition efficiency may be realized byconnecting sub-circuit DSN (FIG. 30A) in parallel with D4. The regulatedboost output PF+ connects to the following: sub-circuit SN pin SNOUT,sub-circuit DSN pin SNOUT and diode D4 cathode. Node PF+ also connectson FIG. 1A to capacitors [C16∥C17∥R17], sub-circuit IPFFB (FIG. 40) pinPF+, sub-circuit PPT1 (FIG. 19) pin P2CT, snubber sub-circuit SNA (FIG.31) pin SNA3, and snubber SNB (FIG. 32) pin SNB3. Magnetic elementwinding pin S1H of sub-circuit PFT1 is connected to CP pin CT1A and pinPFVC of sub-circuit PFA. Magnetic element winding node S1L ofsub-circuit PFT1 is connected to CP pin CT2A. Magnetic element windingnode S2H of sub-circuit PFT1 is connected to pin 10 FIG. 1A then to CPApin CT1B. Magnetic element winding node S2L of sub-circuit PFT1 isconnected to pin 12 FIG. 1A then to CPA pin CT2B. Sub-circuit PFA usingthe AC line phase, load voltage, and magnetic element feedback,generates a command pulse PFCLK. Pin PFCLK of sub-circuit PFA (FIG. 23)is connected to the input of buffer amplifier pin GA1 of sub-circuitAMP1 (FIG. 29). Buffered high-speed gate drive output pin GA2 ofsub-circuit AMP is connected to gate of switch FET Q1. The bufferingprovided by AMP shortens switch Q1 ON and OFF times greatly reducingswitch losses (see FIG. 13 & 14). The source of Q1 with pin GA0 isconnected to return node BR−. Power to sub-circuit AMP is connected topin GA+ from sub-circuit OTP pin TS+. Thermal switch THS1 is connectedto Q1. In the event the case of Q1 reaches approximately 105 C. THS1opens removing power to sub-circuit AMP, safely shutting down the first(input) stage. Normal operation resumes after the switch temperaturedrops 20-30 deg. C. closing THS1. Drain of output switch Q1 is connectedto primary winding pin P1A of non-saturating magnetic sub-circuit PFT1(FIG. 18) and to pin SNL2 of snubber sub-circuit SN (FIG. 30). Referencevoltage from PFC sub-circuit PFA pin PFA2 is connected to feedbacknetworks sub-circuit IPFFB pin FBC and to sub-circuit IFB pin FBC.Control current feedback networks is summed at node PF1 of sub-circuitPFA. Pin PF1 is connected to feed back networks sub-circuit IPFFB pinFBE and to sub-circuit IFB pin FBE. Constant frequency/duty-cyclenon-overlapping two-phase generator sub-circuit PPG (FIG. 43 1A)generates the drive for the push-pull output stage. Phase one output pinPH1 is connected to sub-circuit AMP1 in GA1, second phase output pin PH2is connected to sub-circuit AMP2 pin GA1. Output of amplifier buffersub-circuit AMP1 pin GAP2 connects to gate of push-pull output switchQ6. Output of amplifier buffer sub-circuit AMP2 pin GAP2 connects togate of push-pull output switch Q9. The buffering currents from AMP1 andAMP2 provide fast, low impedance critically damped switching to Q6 andQ9 greatly reducing ON-OFF transition time and switching losses.Regulated 18-volt power from sub-circuit CPA (FIG. 1A) pin CP2+ isconnected to amplifier buffer sub-circuit AMP1 pin GA+, amplifier buffersub-circuit AMP2 pin GA+ and sub-circuit PPG pin PPG+. Drain oftransistor Q6 is connected to snubber network sub-circuit SNB pin SNB1and to non-saturating center tapped primary magnetic element sub-circuitPPT1 pin P2H. Drain of transistor Q9 is connected to snubber networksub-circuit SNA (FIG. 31) pin SNA1 and sub-circuit PPT1 pin P2L. Sourceof transistor Q6 is connected to snubber network sub-circuit SNB pinSNB2, transistor Q9 source, sub-circuit SNA pin SNA2 and to return nodeBR+. Isolated output of NSME sub-circuit PPT1 pin SH connects to Pin C7Bof rectifier sub-circuit OUTA (FIG. 25A), pin SL connects to sub-circuitOUTA C8B. Center tap of PPT1 pin SCT is the output return or negativenode OUT− it connects to sub-circuit OUTA pin OUT− and sub-circuit IFB(FIG. 40B) pin OUT− and RLOAD. Converter positive output fromsub-circuit OUTA pin OUT+ is connected to RLOAD and sub-circuit IFB pinOUT+. FIG. 1 elements LL1, BR, PFA, AMP, Q1, IPFFB, IFB and PFT1 (inputstage) perform power factor corrected AC to DC conversion. The regulatedhigh voltage output of this converter supplies the efficient fixedfrequency/duty-cycle push-pull stage comprising PPG, AMP1, AMP2, Q6, Q9,PPT1 and OUTA (FIG. 1A). Magnetic element sub-circuit PPT1 providesgalvanic isolation and minimal voltage overshoot and ripple in thesecondary thus minimizing filtering requirements of the rectifiersub-circuit OUTA. Five volt reference output from sub-circuit PFA pinPFA2 connects to pin 15 then to FIG. 1A sub-circuit IPFFB pin FBC and tosub-circuit IFB pin FBC. Pulse width control input from sub-circuit PFApin PF1 connects to pin 14 then to FIG. 1A sub-circuit IPFFB pin FBE andto sub-circuit IFB pin FBE. Sub-circuit IFB provides high-speed feedbackto the AC DC converter, the speed of the boost stage provides preciseoutput voltage regulation and active ripple rejection. In the event ofsudden line or load changes, sub-circuit IPFFB corrects the internalboost to maintain regulation at the isolated output. Remote load sensingand other feedback schemes known in the art may be implemented withsub-circuit IPFFB. This configuration provides power factor correctedinput transient protection, rapid line-load response, excellentregulation, isolated output and quiet efficient operation at hightemperatures.

[0090]FIG. 2 is a schematic diagram of an embodiment of a DC to ACconverter. The invention DCAC1 is an efficient push-pull converter.Comprised of sub-circuits PPG (FIG. 43), AMP1 (FIG. 29), AMP2 (FIG. 29),SNB (FIG. 32), SNA (FIG. 31), PPT1 (FIG. 19) and OUTA (FIG. 25),switches Q6 and Q9. Table Element Value/part number Q6 FS 14SM-18 A Q9FS 14SM-18 A

[0091] Converter ACDC1 accepts variable DC voltage and efficientlyconverts it to a variable AC voltage output at a fixed frequency.Variable frequency operation may be achieved by simple changes to PPG.In this embodiment fixed frequency operation is required. The magneticelement comprises non-saturating magnetics. A graph of B-H hysteresisfor the non-saturating magnetics is set forth in FIG. 15. Variable DCvoltage is applied to pin DC+. The pin DC+ connects to the following,sub-circuit PPT1 (FIG. 19) pin P2CT, snubber sub-circuit SNA (FIG. 31)pin SNA3, and snubber SNB (FIG. 32) pin SNB3. Constant frequencynon-overlapping two-phase generator sub-circuit PPG (FIG. 43) generatesthe drive for the push-pull output switches. Phase one output pin PH1 isconnected to sub-circuit AMP1 pin GA1, the second phase output pin PH2is connected to sub-circuit AMP2 pin GA1. Output of amplifier buffersub-circuit AMP1 pin GAP2 connects to gate of push-pull output switchQ6. Output of amplifier buffer sub-circuit AMP2 pin GAP2 connects togate of push-pull output switch Q9. The buffering provided by AMP1 andAMP2 shortens switch Q1 ON and OFF times greatly reducing switchinglosses (See FIG. 13 and 14). External regulated 18-volt power from pinP18V connected to amplifier buffer sub-circuit AMP1 pin GA+, amplifierbuffer sub-circuit AMP2 pin GA+ and sub-circuit PPG pin PPG+. Drain oftransistor Q6 is connected to snubber network sub-circuit SNB pin SNB1and to non-saturating center tapped primary magnetic element sub-circuitPPT1 pin P2H. Drain of transistor Q9 is connected to snubber networksub-circuit SNA (FIG. 31) pin SNA1 and sub-circuit PPT1 pin P2L. Sourceof transistor Q6 is connected to snubber network sub-circuit SNB pinSNB2, transistor Q9 source, sub-circuit SNA pin SNA2, sub-circuit AMP1pin GA0, sub-circuit AMP2 pin GA0, sub-circuit PPG pin PPG0, and toreturn pin DC−. AC output of NSME sub-circuit PPT1 pin SH connects toPin ACH, pin SL connects to pin ACL. Center tap of PPT1 pin SCT isconnected to pin AC0. Magnetic element sub-circuit PPT1 providesgalvanic isolation and minimal voltage overshoot in the secondary thusminimizing filtering requirements if a rectifier assembly is attached.Sub-circuit DCAC1 may be used as a stand-alone converter or as a fastquiet efficient stage in a multi stage converter system. Sub-circuitDCAC1 achieves isolated output, quiet operation, efficient conversion,and operation at high and low temperatures.

[0092]FIG. 3 and 3A is a three-stage version of the present invention.The arrangement is comprised of an AC-DC or DC-DC boost converter stage,DC-DC forward converter stage, and a push-pull stage. This systemreduces losses by combining low current buck regulation, bufferedswitching, rectified snubbering, and NSME in each stage. A power factorcorrected boost stage is used to assure that any load connected to theconverter looks like a resistive load to the AC line, eliminatingundesirable harmonic and displacement currents in the AC power line.NSME having a lower permeability compared to the prior art are used tominimize magnetizing losses, improve coupling efficiency, minimizemagnetic element heating, eliminate saturated core current spikes/gapleakage, reduce parts count, reduce thermal deterioration, and increaseMTBF (mean time before failure). The invention also uses an emitterfollower circuit with a high speed switching FET to slew the main FETgate rapidly. The use of non-saturating magnetics allows operation athigher voltages, which proportionally lowers current further reducingswitch, magnetic element, and conductor losses due to I²R heating. Highvoltage FET switches have the added benefit of lower gate capacitance,which translates to faster switching. At turn on, the n-channel gatedrive FET quickly charges the main FET gate. At turn off, a PNPDarlington transistor switch quickly discharges the main FET gate. Theflyback effect in the PFC stage is managed by use of rectifying RCnetworks positioned across the output diode with an additional capacitorcoupled diode across the switched magnetic element to decouple andfurther dampen the inductive flyback.

[0093]FIG. 3 and FIG. 3A is a schematic diagram of a three stage AC toDC converter. FIG. 3 and 3A is a three-stage version of the presentinvention. The arrangement is comprised of an AC-DC or DC-DC boostconverter stage, DC-DC forward converter stage, and a push-pull stage.This system reduces losses by combining low current buck regulation,buffered switching, rectified snubbering, and NSME in each stage. Apower factor corrected boost stage is used to assure that any loadconnected to the converter looks like a resistive load to the AC line,eliminating undesirable harmonic and displacement currents in the ACpower line. NSME having a lower permeability compared to the prior artare used to minimize magnetizing losses, improve coupling efficiency,minimize magnetic element heating, eliminate saturated core currentspikes/gap leakage, reduce parts count, reduce thermal deterioration,and increase MTBF (mean time before failure). The invention also uses anemitter follower circuit with a high speed switching FET to slew themain FET gate rapidly. The use of non-saturating magnetics allowsoperation at higher voltages, which proportionally lowers currentfurther reducing switch, magnetic element, and conductor losses due toI²R heating. High voltage FET switches have the added benefit of lowergate capacitance, which translates to faster switching. At turn on, then-channel gate drive FET quickly charges the main FET gate. At turn off,a PNP Darlington transistor switch quickly discharges the main FET gate.The flyback effect in the PFC stage is managed by use of rectifying RCnetworks positioned across the output diode with an additional capacitorcoupled diode across the switched magnetic element to decouple andfurther dampen the inductive flyback. The invention is comprised of apower factor corrected regulating boost stage with line protectionfilter sub-circuit LL1 (FIG. 21) and full-wave rectifier sub-circuit BR(FIG. 22) and capacitors C1 and C2. Sub-circuits PFB (FIG. 24), resistorR2, rectifier CP (FIG. 26), magnetic element PFT1 (FIG. 18), overtemperature protection OTP (FIG. 28) snubber SN (FIG. 30) gate bufferAMP (FIG. 29), switch transistors Q1, flyback diode D4, holdupcapacitors C17 and C16, bleed resistor R17, and voltage feedbacksub-circuit FBA (FIG. 40A). An efficient second pre-regulating buckstage with sub-circuits PWFM (FIG. 33), current sense resistor R26,rectifier CPA (FIG. 27), magnetic element BL1, (FIG. 18B), over voltageprotection OVP (FIG. 42), IPFFB (FIG. 40) gate buffer AMP3 (FIG. 29),switch transistor Q2, flyback diode D70, storage capacitor C4, andvoltage feedback sub-circuit IFB (FIG. 40B).

[0094] An efficient third push-pull isolation stage with sub-circuitsCPA (FIG. 27), two-phase generator PPG (FIG. 43), gate buffers AMP1(FIG. 29) and AMP2 (FIG. 29), switch transistors Q6, and Q9, snubbersSNA (FIG. 31) and SNB (FIG. 32), magnetic element PPT1 (FIG. 19) andrectifier OUTA (FIG. 25). FIG. 3 3A Table Element Value/part number C1.01 uf C2 1.8 uf R2 100k ohms D4 STA1206 DI R17 375k ohms Q1 IRFP460 C16100 uf C17 100 uf R26 .05 ohms D70 STA1206 DI Q2 IRFP460 C4 10 uf Q6FS14Sm-18 A Q9 FS14Sm-18 A

[0095] AC line is connected to sub-circuit LLA (FIG. 21) between pinsLL1 and LL2. AC/earth ground is connected to node LL0. The filtered andvoltage limited AC line appears on node/pin LL5 of sub-circuit LLA andconnected to node BR1 of bridge rectifier sub-circuit BR. The neutral/ACreturn leg of the filtered and voltage limited AC appears on pin LL6 ofsub-circuit LL is connected to input pin BR2 of BR. The line voltage isfull-wave rectified and is converted to a positive haversine appearingon node BR+ of sub-circuit BR. Start up resistor R2 connects BR+ tosub-circuit CP pin CP+. Node CP+ connects to pins PFA+ of controlelement sub-circuit PFB and over temperature switch sub-circuit OTP pinGAP. Resistor R2 provides start up power to the control element untilthe regulator CP is at full output. Node S1H from PFT1 is connected topin 31 (FIG. 3) then to pin CT1A of sub-circuit CP and pin PFVC ofsub-circuit PFB. The zero crossing of the core bias are sensed when thevoltage at S1H is at zero relative to BR−. The core zero crossings areused to reset the PFC and start a new cycle. The positive node of the DCside of bridge BR+ is connected through capacitor C2 to BR−. CapacitorC2 is selected for various line and load conditions to de-coupleswitching current from the line improving power factor. Sub-circuit BRpin BR+ connects to pin SNL1 of snubber sub-circuit SN, sub-circuit PFBpin BR+ and pin BR+ (FIG. 3A) then to primary of NSME sub-circuit PFT1pin P1B and to sub-circuit OVP pin BR+. The return line for therectified AC power is connected to the following pins; BR− ofsub-circuit BR, sub-circuit PFT1 pin S1CT, PFC sub-circuit PFB pin BR−,sub-circuit FBA pin BR−, capacitor C2, sub-circuit CP pin CT0,sub-circuit IPFFB pin FBE, and through EMI filter capacitor C1 to earthground node LL0. Node BR− continues to FIG. 3A connecting to R26,capacitors [C16∥C17∥R17], sub-circuit OVP pin BR−, sub-circuit PWFM pinPWFM0, sub-circuit AMP3 pin GA0, switch Q2 source. Floating ground nodePF− is connected to magnetic element sub-circuit PFT1 pin S2CT,rectifier sub-circuit CPA pin CT20, generator sub-circuit PPG (FIG. 43)pin PPG0, sub-circuit AMP1 pin GA0, sub-circuit AMP2 pin GA0, capacitorC4, magnetic element BL1, pin, transistor Q6 source, transistor Q9source, sub-circuit SNA pin SNA2 sub-circuit SNB pin SNB2, pin PF-FIG. 3then to sub-circuit IPFFB pin PF−. Drain of output switch Q1 isconnected to diode D4 anode, sub-circuit SN pin SNL2, then to pin 34 ofFIG. 3A then to sub-circuit PFT1 pin P1A. Snubber SN reduces the highvoltage stress to Q1 until flyback diode D4 begins conduction.Additional rectification efficiency and protection is achieved by addingsub-circuit DSN (FIG. 30A) across flyback diode D4. Feedback correctedboost output voltage of the power factor corrected AC to DC converterstage appears across nodes PF+ and PF−. The regulated 385-volt boostoutput node PF+ connects to the following; sub-circuit SN pin SNOUT,diode D4 cathode, sub-circuit IPFFB (FIG. 40) pin PF+, sub-circuit FBApin PF+, then to pin PF+ of FIG. 3A, capacitors [C16∥C17∥R17], magneticelement sub-circuit PTT1 (FIG. 19) pin P2CT, snubber sub-circuit SNA(FIG. 31) pin SNA3, and snubber SNB (FIG. 32) pin SNB3, sub-circuit OVPpin PF+, capacitor C4 and diode D70 cathode. Magnetic element windingnode S1H of sub-circuit PFT1 is connected to pin 31 FIG. 3 then tosub-circuit CP pin CT1A and pin PFVC of sub-circuit PFB. Magneticelement winding node S1L of sub-circuit PFT1 is connected to pin 33 FIG.3 then to sub-circuit CP pin CT2A. Magnetic element winding node S2H ofsub-circuit PFT1 is connected to CPA pin CT1B. Magnetic element windingnode S2L of sub-circuit PFT1 is connected to CP pin CT2B. Sub-circuitPFB using feedback from the phase of the AC line, Q1 switch current,magnetic bias first stage and output voltage feedback generates acommand pulse on pin PFCLK. Pin PFCLK of sub-circuit PFB (FIG. 24) isconnected to the input of buffer AMP amplifier pin GA1 of sub-circuitAMP1. Buffered high-speed low impedance gate drive output pin GA2 ofsub-circuit AMP is connected to gate of switch FET Q1. The bufferingprovided by AMP shortens switch Q1 “ON” and “OFF” times greatly reducingswitch losses (See FIGS. 13 and 14). The source of Q1 is connected tosub-circuit AMP pin GA0, pin 35 of FIG. 3A then to current senseresistor R26 connected to return node BR−. The voltage developed acrossR26 is fed back to PFB pin PFSC. This signal is used to protect theswitch by reducing the pulse width in response to a low line or highload induced over current fault. The return line of sub-circuit FBA pinBR− is connected to node BR− and to pin BR− of sub-circuit PFB. Thisfeedback is non-isolated; network values are selected for the firststage to develop a 385-Volt output at PF+. Sub-circuit feedback networkFBA (FIG. 40A) pin PF1 is connected to sub-circuit PFB pin PF1.Controller PFB modulates PFCLK signal to maintain a substantiallyconstant 385-voltage at PF+ independent of line and load conditions. Inthe event of a component failure in sub-circuit FBA the PBF may commandthe converter to very high voltages. Sub-circuit OVP monitors the firststage boost in the event it exceeds 405-volts OVP will clamp the outputof sub-circuit BR causing fuse F1 in sub-circuit LLA to open. Analternate over voltage network OVP1 (FIG. 42A) may replace OVP clampingthe 18-volt control power stopping the boost action of the converterwithout opening the fuse. Sampled converter output at node fromsub-circuit FBA pin PF1 is connected to sub-circuit PFB pin PF1. Thehaversine on BR+ is used with an internal multiplier by PFB to generatevariable width control pulses on pin PFCLK. The high frequencymodulation of switch Q1 makes the load/converter appear resistive to theAC line. Over temperature protection sub-circuit OTP pin TS+ isconnected to sub-circuit AMP pin GA+. Thermal switch THS1 is connectedto Q1. In the event Q1 reaches approximately 105 C. THS1 opens removingpower to sub-circuit AMP, safely shutting down the first stage. Normaloperation resumes after the temperature decreases 20-30 C. closing THS1.The second stage is configured as a buck stage. It accepts the 385-Voltoutput of the first stage. By employing a second floating reference nodePF− energy storage element capacitor C4 the voltage to the finalpush-pull stage may be regulated with minimal loss. Power fromsub-circuit CP pin CP18V+ is connected to pin 30 of FIG. 3A then tosub-circuit PWFM (FIG. 33) pin PWM+ and AMP3 pin GA+. Feedback currentfrom sub-circuit IPFFB pin FBC is connected to pin 36 FIG. 3A then tosub-circuit IFB pin FBC and sub-circuit PWFM pin PF1. Sub-circuit IPFFBonly shunts current from this node if the output of the second stage isgreater than 200-volts. When the converter reaches its designed outputvoltage, IFB shunts current from PWFM pin PF1 signaling PWFM to reducethe pulse width on pin PWMCLK. Sub-circuit AMP3 input pin is connectedto sub-circuit PWFM pin PWMCLK. Output of AMP3 buffer pin GA2 isconnected to gate of switch Q2. Drain of Q2 is connected to anode of D70and non-saturating magnetic sub-circuit BL1, pin P2B (FIG. 18B). Turningon switch Q2 charges C4 also storing energy in magnetic element BL1.Releasing switch Q2 allows energy stored in magnetic element BL1, tocharge C4 through flyback diode D70. Larger pulse widths charge C4 tolarger voltages thus efficiently blocking part of the first stagevoltage to the final push-pull stage. This action provides regulatedvoltage to the final converter stage. The third and final push-pull(transformer) converter stage provides the galvanic isolation, filteringand typically converts the internal high voltage bus to a lowerregulated output voltage. The efficient push-pull stage producesalternating magnetizing currents in the NSME for maximum load over coremass. Constant frequency non-overlapping two-phase generator sub-circuitPPG (FIG. 43) generates the drive for the push-pull output stage. Phaseone output pin PH1 is connected to sub-circuit AMP1 pin GA1, output pinPH2 is connected to sub-circuit AMP2 pin GA1. Output of amplifier buffersub-circuit AMP1 pin GAP2 connects to gate of push-pull output switchQ6. Output of amplifier buffer sub-circuit AMP2 pin GAP2 connects togate of push-pull output switch Q9. The buffering provided by AMP1 andAMP2 shortens switch Q1 ON and OFF times greatly reducing switchinglosses. (See FIG. 13 and 14) Regulated 18-volt power from sub-circuitCPA pin CP18+ is connected to amplifier buffer sub-circuit AMP1 pin GA+,amplifier buffer sub-circuit AMP2 pin GA+ and sub-circuit PPG pin PPG+.Drain of transistor Q6 is connected to snubber network sub-circuit SNBpin SNB1 and to non-saturating center tapped primary magnetic elementsub-circuit PPT1 pin P2H. Drain of transistor Q9 is connected to snubbernetwork sub-circuit SNA (FIG. 31) pin SNAL and sub-circuit PPT1 pin P2L.Return node PF− connects source of transistor Q6 to snubber networksub-circuit SNB pin SNB3, transistor Q9 source, sub-circuit SNA pin SNA3and to return node GND2. Output of NSME sub-circuit PPT1 pin SH connectsto pin C7B of rectifier sub-circuit OUTA (FIG. 25), pin SL connects toC8B. Center tap of PPT1 pin SCT is the output return or negative nodeOUT− it connects to sub-circuit pin OUT− and sub-circuit IFB pin OUT−and RLOAD. Supply positive output from sub-circuit OUTA pin OUT+ isconnected to RLOAD and sub-circuit IFB pin OUT+. Elements LL1, BR, PFA,AMP, Q1, IPFFB, IFB and PFT1 provide power factor corrected AC to DCconversion and DC output regulation. The regulated high voltage outputof this converter is used to power the efficient fixed frequencypush-pull stages PPG, AMP1, AMP2, Q6, Q9, PPT1 and OUTA. Magneticelement sub-circuit PPT1 provides galvanic isolation and minimal voltageovershoot in the secondary thus minimizing filtering requirements of therectifier sub-circuit OUTA. Sub-circuit IFB provides high-speed feedbackto the AC DC converter, the speed of the boost stage provides preciseoutput voltage regulation and active ripple rejection. In the event of asudden line or load changes sub-circuit IPFFB compensates the internalboost. This system reduces losses by focusing output control in themiddle (low current) stage of the converter and by using non-saturatingmagnetics, buffered switching, and rectifying snubbers throughout eachstage. The combined improvements translate to higher systemefficiencies, higher power densities, lower operating temperatures, and,improved thermal tolerance thereby reducing or eliminating the need forforced air-cooling per unit output. The non-saturating magneticproperties are relatively insensitive to temperature (see FIG. 17), thusallowing the converter to operate over a greater temperature range. Inpractice, the operating temperature for the Kool Mu NSME is limited to200 C. by wire/core insulation; the non-saturating magnetic materialremains operable to near its Curie temperature of 500 C. Thisconfiguration provides power factor corrected input transientprotection, rapid line-load and ripple compensation, excellent outputregulation, output isolation and quiet efficient operation at hightemperatures.

[0096]FIG. 4 is a schematic diagram sub-circuit ACDCPF.

[0097]FIG. 4 is a schematic diagram of a power factor corrected singlestage AC to DC converter sub-circuit ACDCPF. The invention is comprisedof line protection filter sub-circuit LL (FIG. 20) and full-waverectifier sub-circuit BR (FIG. 22). A power factor corrected regulatedboost stage with sub-circuits PFB (FIG. 24), snubber sub-circuit SN(FIG. 30), magnetic element sub-circuit PFT1A (FIG. 18A), sub-circuit CP(FIG. 26), buffer sub-circuit AMP (FIG. 29), over temperaturesub-circuit OTP (FIG. 28), and voltage feedback sub-circuit FBA (FIG.40A). Start up resistor R2, filter capacitor C1, PFC capacitor C2,flyback diode D4, switch transistor Q1, hold up capacitors C17 and C16,and resistor R17. Table Element Value/part number Cl .01 uf C2 1.8 uf R2100k ohms R26 0.05 ohms Q1 IRFP 460 D4 STA1206 DI C17 100 uf C16 100 ufR17 375k ohms

[0098] AC line is connected to sub-circuit LL (FIG. 20) between pins LL1and LL2. AC/earth ground is connected to node LL0. The filtered andvoltage limited AC line appears on node/pin LL5 of sub-circuit LL1 andconnected to node BR1 of bridge rectifier sub-circuit BR (FIG. 22). Theneutral/AC return leg of the filtered and voltage limited AC appears onpin LL6 of sub-circuit LL is connected to input pin BR2 of BR. The linevoltage is full-wave rectified and is converted to a positive haversineappearing on node BR+ of sub-circuit BR (FIG. 22). Start up resistor R2connects BR+ to sub-circuit CP pin CP+. Node CP+ connects to pins PFA+of power factor controller sub-circuit PFA (FIG. 24) and overtemperature switch sub-circuit OTP (FIG. 28) pin GAP. Resistor R2provides start up power to the control element until the rectifier andregulator CP is at full output. Node S1H from PFT1A is connected to nodePFVC sub-circuit PFB. The zero crossing of the core bias are sensed whenthe voltage at S1H is at zero. The core zero crossings are used to resetthe PFC and start a new cycle. The positive node of the DC side ofbridge BR+ is connected through capacitor C2 to BR−. C2 is selected forvarious line and load conditions to de-couple switching current from theline improving power factor. Primary of NSME sub-circuit PFT1A (FIG.18A) pin P1B connects to pin SNL1 of snubber sub-circuit SN (FIG. 30),sub-circuit PFB pin BR+ and connects to node BR+. The return line forthe rectified AC power BR− is connected to the following pins; BR− ofsub-circuit BR, sub-circuit PFB pin BR−, sub-circuit AMP pin GA0, senseresistor R26, capacitor [C16∥C17∥resistor R17], capacitor C2,sub-circuit CP pin CT0, sub-circuit PFT1A pin S1CT and through EMIfilter capacitor C1 to earth ground node LL0. Drain of output switch Q1is connected to diode D4 anode, sub-circuit PFT1A pin P1A and snubbersub-circuit SN pin SNL2. Additional rectification efficiency andprotection is achieved by adding sub-circuit DSN (FIG. 30A) in parallelflyback diode D4. Sub-circuit provides reduces the high voltage stressto Q1 until flyback diode D4 begins conduction. Line coupled, powerfactor corrected boost regulated output voltage of the AC to DCconverter stage (FIG. 1) appears on node PF+. The regulated boost outputPF+ connects to the following; sub-circuit SN pin SNOUT, diode D4cathode, capacitor [C16∥C17∥R17], and snubber DSN (FIG. 30A) pin SNOUT.Magnetic element winding node S1H of sub-circuit PFT1A is connected toCP pin CT1A and pin PFVC of sub-circuit PFB. Magnetic element windingnode S1L of sub-circuit PFT1A is connected to CP pin CT2A. Sub-circuitPFB using the phase of the AC line, and load voltage generates a commandpulse PFCLK. Pin PFCLK of sub-circuit PFB (FIG. 24) is connected to theinput of buffer amplifier pin GA1 of sub-circuit AMP1 (FIG. 29).Buffered high-speed gate drive output pin GA2 of sub-circuit AMP isconnected to gate of switch FET Q1. The buffering provided by AMPshortens switch Q1 ON and OFF times greatly reducing switch losses. Thesource of Q1 is connected to current sense resistor R26, pin PFSC ofsub-circuit PFB, connected then to return node BR−. The voltagedeveloped across R26 is feedback to PFB pin PFSC. This signal is used toprotect the switch in the event of an over current fault. Thermal switchTHS1 is connected to Q1. In the event Q1 reaches approximately 105 C.THS1 opens removing power to sub-circuit AMP, safely shutting down thefirst stage. Normal operation resumes after the switch temperature drops20-30 C. closing THS1. Sub-circuit feedback network FBA (FIG. 40A) pinPF1 is connected to sub-circuit PFB pin PF1. Converter output at nodePF+ (the junction of C17∥C16 and D4) is connected to sub-circuit FBA pinPF+. The return line of sub-circuit FBA pin BR− is connected to pin BR−of sub-circuit PFB. This feed back is non-isolated; network values areselected for a substantially constant 385-Volt output at PF+ relative toBR−. The high-voltage haversine from the rectifier section BR pin BR+ isconnected to sub-circuit PFB pin BR+. The haversine is used with aninternal multiplier by PFB to make the converter ACDCPF appear resistiveto the AC line. Sub-circuits LL1, BR, PFB, AMP, Q1, OTP, FBA, IFB andPFT1A perform power factor corrected AC to DC conversion. The regulatedhigh voltage output of this converter may be used use to power one ormore external converters connected to the PF+ and BR− nodes. The NSMEsub-circuit PPT1A provides efficient boost action at high power levelsin a very small form factor. Sub-circuit FBA provides high-speedfeedback to the converter the speed of the boost stage provides preciseoutput voltage regulation and active ripple rejection. Thisconfiguration provides power factor corrected input transientprotection, rapid line-load response, excellent regulation, and quietefficient operation at high temperatures.

[0099]FIG. 5 is a graph comparing typical currents in saturating andnon-saturating magnetic elements. As the inductance does not radicallychange at high temperatures and currents in the NSME, the large currentspikes due to the rapid reduction of inductance common in saturatingmagnetics is not seen. As a result, destructive current levels,excessive gap leakage, magnetizing losses, and magnetic element heatingare avoided in NSME.

[0100]FIG. 6 is a schematic for non-isolated low side switch buckconverter sub-circuit NILBK. Sub-circuit NILBK consists of resistor R20,diode D6, capacitor C6, FET transistor Q111, sub-circuit CP (FIG. 26),sub-circuit PFT1A (FIG. 18A), sub-circuit IFB (FIG. 40B), sub-circuitAMP (FIG. 29) and sub-circuit PWFM (FIG. 33). Table Element Value/partnumber R20 100k ohms R20 STA1206 DI Q111 IRFP460 C6 10 uf

[0101] External power source VBAT connects to pins DCIN+ and DCIN−. FromDCIN+ through resistor R20 connects to sub-circuit CP pin CP+,sub-circuit AMP pin GA+and to sub-circuit PWFM pin PWFM+. Resistor R20provides startup power to the converter before regulator sub-circuit CPreaches it full 18-volt output. VBAT negative is connected to pin DCIN−connects to sub-circuit PWFM pin PWFM0, sub-circuit AMP pin GA0, Q111source, sub-circuit IFB pin FBE, sub-circuit CP pin CT0, and sub-circuitPFT1 pin SICT. Magnetic element winding node S1H of sub-circuit PFT1A isconnected to CP pin CT1A. Magnetic element winding node S1CT ofsub-circuit PFT1 is connected to CP pin CT0. Magnetic element windingnode S1H of sub-circuit PFT1A is connected to CP pin CT2A. The regulated18 volts from sub-circuit CP+ is connected to R20, sub-circuit AMP pinGA+ and to sub-circuit PWFM pin PWFM+. Sub-circuit PWFM is designed forvariable pulse width operation. PWFM is configured for maximum pulsewidth 90-95% with no feedback current from sub-circuit IFB pin FBC.Increasing the feedback current reduces the pulse-width and outputvoltage from converter NILBK. Sub-circuit PWFM clock/PWM output pin CLKis connected to the input pin GA1 of buffer sub-circuit AMP. The outputof sub-circuit AMP pin GA2 is connected to the gate of Q111. Input nodeDCIN+ connects to the cathode of flyback diode D6, sub-circuit IFB pinOUT+, resistor RLOAD, capacitor C6 and pin B+. The drain of Q111 isconnected to sub-circuit PFT1 pin P1B and the anode of D6. Pin P1A ofsub-circuit PFT1A is connected to capacitor C6, RLOAD, sub-circuit IFBpin OUT− and to node B− With sub-circuit PWFM pin CLK high buffer AMPoutput pin GA2 charges the gate of transistor switch Q111. Switch Q111conducts charging capacitor C10 through NSME PFT1A from source VBAT andstoring energy in PFT1A. Feedback output pin FBC from sub-circuit IFB isconnected to sub-circuit PWFM pulse-width adjustment pin PW1.Sub-circuit IFB removes current from PW1 commanding PWFM to reduce thepulse-width or on time of signal CLK. After sub-circuit PWFM reaches thecommanded pulse-width PWFM switches output pin CLK low turning “off”Q111 stopping the current into PFT1A. The energy not transferred intoregulator sub-circuit CP load is released from NSME PFT1A into the nowforward biased diode D6 charging capacitor C6. By modulating the “on”time of switch Q111 the converter buck voltage is regulated. Regulatedvoltage is developed across Nodes B− and B+. Sub-circuit IFB providesthe isolated feedback voltage to the sub-circuit PWFM. When sub-circuitIFB senses the converter output (nodes B+ and B−) is at the designedvoltage, current from REF is removed from PM1. Sinking current from PM1commands the PWFM to a shorter pulse-width thus reducing the converteroutput voltage. In the event the feedback signal from IFB commands thePWFM to minimum output. Gate drive to switch Q111 is removed stoppingall buck activity capacitor C6 discharges through RLOAD. Input currentfrom VBAT is sinusoidal making the converter very quiet. In addition theswitch Q111 is not exposed to large flyback voltage. Placing less stresson the switches thereby increasing the MTBF. Sub-circuit NILBK takesadvantage of the desirable properties of the NSME in this convertertopology. Adjusting the NSME 100 (FIG. 18A) primary inductance andcomponent values in sub-circuit IFB determines the output buck voltage.

[0102]FIG. 8 is a schematic for a tank coupled single stage convertersub-circuit TCTP. Sub-circuit TCTP consists of resistor R20 and RLOAD,capacitor C10, Darlington transistors Q10 and Q20, sub-circuit CP (FIG.26), sub-circuit PFT1 (FIG. 18), sub-circuit OUTB (FIG. 25A),sub-circuit IFB (FIG. 40B) and sub-circuit PWFM (FIG. 33). Table ElementValue/part number R20 5k ohms Q10 TST541 Q20 IRFP460 C10 1.8 uf

[0103] External power source VBAT connects to pins DCIN+ and DCIN−. FromDCIN+ connects to Q10 collector then through resistor R20 connects tosub-circuit CP pin CP+ and to sub-circuit PWFM pin PWFM+. Resistor R20provides startup power to the converter before regulator sub-circuit CPreaches it full 18-volt output. VBAT negative is connected to pin DCIN−ground/return node GND. Node GND connects to sub-circuit PWFM0 pinPWFM0, Q20 collector, C10, sub-circuit CP pin CT0 and sub-circuit PFT1pin S1CT. Magnetic element winding node S1H of sub-circuit PFT1 isconnected to CP CT1A. Magnetic element winding node S1L of sub-circuitPFT1 is connected to CP CT2A. Magnetic element winding node SICT ofsub-circuit PFT1 is connected to CP pin CT0. Magnetic element windingnode S2H of sub-circuit PFT1 is connected to CP pin CT2A. The regulated18 volts from sub-circuit CP+ is connected to R20 and to sub-circuitPWFM pin PWFM+. Sub-circuit PWFM is designed for a constant 50% dutycycle variable frequency generator. Sub-circuit PWFM clock output pinCLK is connected to the base of Q10 and Q20. The emitters of Q10 and Q20are connected to sub-circuit PFT1 pin P1B. This forms an emitterfollower configuration. Pin P1A of sub-circuit PFT1 is connected throughtank capacitor C10 to node GND. With PWFM CLK pin high forward biasedtransistor Q10 supplies current to the tank from BAT1 charging capacitorC10 through NSME PFT1 and transferring energy into PFT1. Sub-circuitPWFM switches CLK low turning “off” Q10 stopping the current into PFT1.Energy not transferred into the load is released from NSME PFT1 into thenow forward biased PNP transistor Q20 back into capacitor C10. Thus anyenergy not used by the secondary loads is transferred back to theprimary tank to be used next cycle. When the switching occurs at theresonant frequency large circulating currents develop in the tank. AlsoC10 is charged and discharged to very large voltages. Oscillograph inFIG. 35 is the actual voltage developed across capacitor C10 with VBATequal to 18 volts. A very large 229-Volts peak to peak was developedacross the nodes P1A and P1A of NSME PFT1. The large primary voltagegenerates large biases in the NSME PFT1 to be flux harvested by thewindings 102 and 103 (FIG. 18) and transferred to a load or rectifiersub-circuit OUTB. Magnetic element winding node S2L of sub-circuit PFT1is connected to OUTB C8 b. Magnetic element winding node S2H ofsub-circuit PFT1 is connected to C7B of sub-circuit OUTB node OUT−. NodeOUT− is connected to RLOAD, pin B− and to sub-circuit IFB pin OUT−.Rectified power is delivered to pin OUT+ of OUTB and is connected toRLOAD, pin B+ and to sub-circuit IFB pin OUT+. Sub-circuit IFB providesthe isolated feedback signal to the sub-circuit PWFM. Frequency controlpin FM1 of sub-circuit PWFM is connected to sub-circuit IFB pin FBE.Internal reference pin REF of sub-circuit PWFM is connected tosub-circuit IFB pin FBC. PWFM is designed to operate at the resonatefrequency of the tank. When sub-circuit IFB senses the converter outputis at the designed voltage, current from REF is injected into FM1.Injecting current into FM1 commands PWFM to a lower frequency. Operatingbelow resonance reduces the amount of energy added to the primary tankthus reducing the converter output voltage. In the event the feedbacksignal from IFB commands the PWFM to 0 Hz all primary activity stops.Input current from VBAT is sinusoidal making the converter very quiet.In addition the switches Q10 and Q20 are never exposed to the largecirculating voltage (FIG. 35). Placing less stress on the switchesthereby increasing the MTBF. Sub-circuit TCTP takes advantage of thedesirable properties of the NSME in this converter topology. Adjustingsecondary turns allows TCTP to generate very large AC or DC outputvoltages as well as low-voltage high current outputs.

[0104]FIG. 9 is a schematic for non-isolated low side switch boostconverter sub-circuit NILSBST. Sub-circuit NILSBST consists of resistorR20 and RLOAD, diode D6, capacitor C6, FET transistor Q111, sub-circuitCP (FIG. 26), sub-circuit PFT1A (FIG. 18A), sub-circuit FBI (FIG. 41),sub-circuit AMP (FIG. 29) and sub-circuit PWFM (FIG. 33). Table ElementValue/part number R20 100k ohms Q111 IRFP460 D6 STA1206 DI C6 200 uf

[0105] External power source VBAT connects to pins DCIN+ and DCIN−. FromDCIN+ Resistor R20 connects to sub-circuit CP pin CP+, sub-circuit AMPpin GA+ and to sub-circuit PWFM pin PWFM+. Resistor R20 provides startuppower to the converter before regulator sub-circuit CP reaches it full18-volt output. VBAT negative is connected to pin DCIN− and groundreturn node GND. Node GND connects to sub-circuit PWFM pin PWFM0,sub-circuit AMP pin GA0, Q111 source, sub-circuit FBA pin BR−,sub-circuit FBA pin FBA, sub-circuit CP pin CT0, capacitor C6, resistorRLOAD, transistor Q111 source, and sub-circuit PFT1 pin SICT. Magneticelement winding node S1H of sub-circuit PFT1A is connected to CP pinCT1A. Magnetic element winding node S1CT of sub-circuit PFT1 isconnected to CP pin CT0. Magnetic element winding node S2H ofsub-circuit PFT1A is connected to CP pin CT2A. The regulated 18 voltsfrom sub-circuit CP+ is connected to R20, sub-circuit AMP pin GA+and tosub-circuit PWFM pin PWFM+. Sub-circuit PWFM is designed for variablepulse width operation. PWFM is configured for maximum pulse width 90-95%(maximum boost voltage) with no feedback current from sub-circuit FBI.Increasing the feedback current reduces the pulse-width reducing theboost voltage and reducing the output from converter NILSBST.Sub-circuit PWFM clock/PWM output pin CLK is connected to the input pinGA1 of buffer sub-circuit AMP. The output of sub-circuit AMP pin GA2 isconnected to the gate of Q111. Input node DCIN+ connects to the NSMEPFT1A pin P1A. The drain of Q11 is connected to sub-circuit PFT1A pinP1B and the anode of D6. Cathode of diode D6 is connected to sub-circuitFBA pin PF+, resistor RLOAD, C6 and pin BK+. With sub-circuit PWFM pinCLK high buffer AMP output pin GA2 charges the gate of transistor switchQ111. Switch Q111 conducts reverse biasing diode D6 capacitor C10 stopscharging through NSME PFT1A from source VBAT. During the time Q111 isconducting, energy is stored in NSME sub-circuit PFT1A. Feedback outputpin FBC from sub-circuit FBI is connected to sub-circuit PWFMpulse-width adjustment pin PW1. Sub-circuit FBI removes current from PW1commanding PWFM to reduce the pulse-width or on time of signal CLK.After sub-circuit PWFM reaches the commanded pulse-width PFFM switchesCLK low turning “off” Q111 stopping the current into PFT1A. The energynot transferred into regulator sub-circuit CP load is released from NSMEPFT1A into the now forward biased diode D6 charging capacitor C6. Bymodulating the “on” time of switch Q111 the converter boost voltage isregulated. Regulated voltage is developed across Nodes B− and B+.Sub-circuit IFB provides the feedback current to the sub-circuit PWFM.When sub-circuit IFB senses the converter output (nodes B+ and B−) is ator greater than the designed voltage, current is removed from PM1.Sinking current from PM1 commands the PWFM to a shorter pulse-width thusreducing the converter output voltage. In the event the feedback signalfrom IFB commands the PWFM to minimum output. Gate drive to switch Q111is removed stopping all boost activity capacitor C6 charges to VBAT.Input current from VBAT is sinusoidal making the converter very quiet.In addition the switch Q111 is not exposed to large flyback voltage.Placing less stress on the switches thereby increasing the MTBF.Sub-circuit NILBK takes advantage of the desirable properties of theNSME in this converter topology. Adjusting the NSME 100 (FIG. 18A)primary inductance and component values in sub-circuit IFB determinesthe output boost voltage.

[0106]FIG. 10 is a schematic for a two stage isolated DC to DC boostcontrolled push-pull converter BSTPP. Sub-circuit BSTPP consists ofdiode D14, capacitor C14, FET transistor Q14, sub-circuit REG (FIG. 36),sub-circuit BL1, (FIG. 18B), sub-circuit IFB (FIG. 40B), sub-circuit AMP(FIG. 29), sub-circuit DCAC1, and sub-circuit PWFM (FIG. 33). Externalpower source VBAT connects to pins DCIN+ and DCIN−. Table ElementValue/part number Q31 IRFP460 D14 STA1206 DI C14 10 uf

[0107] From pin DCIN+ connects to sub-circuit REG pin RIN+ andsub-circuit BL1, pin P1A. Voltage regulator sub-circuit output pin +18Vconnects to sub-circuit AMP pin GA+ and to sub-circuit PWFM pin PWFM+.Sub-circuit REG provides regulated low voltage power to the controllerand to the main switch buffers. VBAT negative is connected to pin DCIN−and ground return node GND. Node GND connects to sub-circuit PWFM pinPWFM0, sub-circuit AMP pin GA0, Q14 source, capacitor C14, sub-circuitIFB pin FBE, sub-circuit REG pin REG0, sub-circuit DCAC1 pin DC−.Sub-circuit PWFM (FIG. 33) is designed for variable pulse widthoperation. The nominal frequency is between 20-600 Khz PWFM isconfigured for maximum pulse width 90% (maximum boost voltage) with nofeedback current from sub-circuit FBI. Increasing the feedback currentreduces the pulse-width reducing the boost voltage and reducing theoutput from converter BSTPP. Sub-circuit PWFM clock/PWM output pin CLKis connected to the input pin GA1 of buffer sub-circuit AMP (FIG. 29).The output of switch speed up buffer sub-circuit AMP pin GA2 isconnected to the gate of Q14. Input node DCIN+ connects to the NSME BL1,pin P1A. The drain of Q14 is connected to sub-circuit BL1, pin P1B andthe anode of D14. Cathode of flyback diode D14 is connected tosub-circuit DCAC1 pin DC+ and C14. With sub-circuit PWFM pin CLK highbuffer AMP output pin GA2 charges the gate of transistor switch Q14.Switch Q14 conducts reverse biasing diode D14 capacitor C14 stopscharging through NSME BL1, from source VBAT. During the time Q14 isconducting, energy is stored in NSME sub-circuit BL1. Feedback outputpin FBC from sub-circuit IFB is connected to sub-circuit PWFMpulse-width adjustment pin PW1. Sub-circuit IFB removes current from PW1commanding PWFM to reduce the pulse-width or “on” time of signal CLK.After sub-circuit PWFM reaches the commanded pulse-width PFFM switchesCLK low turning “off” Q14 stopping the current into BL1. The energy isreleased from NSME BL1 into the now forward biased flyback diode D14charging capacitor C14. By modulating the “on” time of switch Q14 theconverter boost voltage is regulated. Regulated voltage is developedacross C14 Nodes DC+ and GND is provided to the isolated constantfrequency push-pull DC to AC converter sub-circuit DCAC1 (FIG. 2).Sub-circuit DCAC1 provides efficient conversion of the regulated boostvoltage to a higher or lower voltage set by the magnetic element-windingratio The center tap of the push-pull output magnetic is connected to,sub-circuit OUTB pin OUT−, RLOAD, sub-circuit IFB pin OUT− and the pinOUT− forming the return line for the load and feedback network. Outputof sub-circuit DCAC1 pin ACH is connected to sub-circuit OUTB pin C7b.Output of sub-circuit DCAC1 pin ACL is connected to sub-circuit OUTB pinC8 b. Sub-circuit OUTB provides rectification of the AC power generatedby sub-circuit DCAC1. Since the non-saturating magnetic converter haslow output ripple, minimal filtering is required by OUTB. This furtherreduces cost and improves efficiency as losses to filter components areminimized. Sub-circuit IFB provides the isolated feedback current to thesub-circuit PWFM. When sub-circuit IFB senses the converter output(nodes OUT+ and OUT−) is greater than the designed/desired voltage,current is removed from node PM1. Sinking current from PM1 commands thePWFM to a shorter pulse-width thus reducing the converter outputvoltage. In the event the feedback signal from IFB commands the PWFM tominimum output. Gate drive to switch Q14 is removed stopping all boostactivity capacitor C14 charges to VBAT. As the non-saturating does notsaturate the destructive noisy current “spikes” common to prior art areabsent. Input current from VBAT to charge C14 is sinusoidal making theconverter very quiet. In addition the switch Q14 is not exposed apotentially destructive current spike. Placing less stress on theswitches thereby increasing the MTBF. Sub-circuit BSTPP takes advantageof the desirable properties of the NSME. Adjusting the NSME BL1, (FIG.18B) sets the amount of boost voltage available to the final push-pullisolation stage. Greater efficiencies are achieved at higher voltages.The final output voltage is set by the feedback set point and the turnsratio of the push-pull element PPT1 (FIG. 19).

[0108]FIG. 11 is a graph of permeability as a function of temperaturefor typical prior art magnetic element material. The high permeabilitymaterial in FIG. 11 exhibits large changes in permeability of almost100% over a 100 C. range as compared to the less than 5% change for thematerial in FIG. 17. The increase in permeability at high temperaturesof the prior art material increases the flux density resulting in coresaturation for a constant power level. (See FIG. 12) Thus the prior artcore must be derated at least 100% to operate over extendedtemperatures. The instant invention takes advantage of the desirableproperties of the NSME. Eliminating the need to derate the magneticelement. As the magnetic element performs better at high temperatures,currently limited by melting wire insulation.

[0109]FIG. 12 is a graph of flux density as a function of temperaturefor typical prior art magnetic element material. The reduction ofmaximum flux density with temperature is typical of the saturatingmagnetic element prior art material. Thus the prior art core is commonlyderated at least 100% to operate over extended temperatures. Resultingin a larger more expensive design, and or the requirement to cool thecore.

[0110]FIG. 12A is a graph of magnetic element losses for various fluxdensities and operating frequencies typical of prior art magneticelement material.

[0111]FIG. 13 is a graph showing standard switching losses. The hashedarea represents the time when the switch is in a resistive state. Thehashed area is proportional to the amount of energy lost each time theoutput switch operates. Total power lost is the product of the loss perswitch times the switching frequency.

[0112]FIG. 14 is a graph showing the inventions switching losses. Thehashed area represents the time when the switch is in a resistive state.The smaller hashed area is due to the action of the buffer in FIG. 29and the snubber isolation diode D805 in FIG. 30. Generally the NSME hasa wider usable frequency band and can be magnetized from higher primaryvoltages. Higher operating voltages have proportionally smaller currentsfor a given power level thus proportionally lower losses. Switchinglosses more closely resemble I²R losses. Most switching loss occursduring turn “on” and turn “off” transitions; total switching losses arereduced proportionally by the lower switching frequencies and fastertransition times characteristic of the disclosed NSME converters. Inaddition the properties of the NSME allow operation at temperatureextremes beyond the tolerance of standard prior art magnetics and theirgeometry's. The combined contributions of the above yields a converterthat requires little or no forced air-cooling. (See FIG. 15, 16, and 17)

[0113]FIG. 15 is a graph of the NSME magnetization curves for Kool Mumaterial. The invention makes advantageous use of the availablesaturation range of the NSME.

[0114]FIG. 16 is a graph of the Kool Mu NSME losses for various fluxdensities and operating frequencies. It can be seen from the data thatmuch higher flux densities are available per unit losses over the priorart.

[0115]FIG. 17 is a plot of permeability vs. temperature for several KoolMu materials. This data demonstrates the usefulness and stability of themagnetic properties over temperature.

[0116]FIG. 18 is a schematic representation of the non-saturatingmagnetic boost element PFT1. Sub-circuit PFT1 consists of a primarywinding 100 around a NSME 101 with two center-tapped windings 102 and103. Table Element Value/part number 100 55 turns 203 uh 101 2 ×77932-A7 102 14 turns 102 14 turns

[0117] The primary winding 100 has nodes P1B and P1A for connections toexternal AC source. Secondary 102 winding has center-tapped node S1CTand node S1H and S1L connections to the upper and lower halvesrespectively. Secondary 103 winding has center-tapped node S2CT and nodeS2H and S2L connections to the upper and lower halves respectively. Both102 and 103 are connected to external full-wave rectifier assemblies.Magnetic element magnetic element 101 comprises a non-saturating, lowpermeability magnetic material. The permeability is on the order of 26 uwith a range of 1 u to 550 u, as compared to the prior art, which rangesfrom 1500 to 5000 u. Flyback management is of concern when using NSME ina boost converter because the magnetic element generates high drainsource voltages across the primary switch during the reverse recoverytime of the flyback (output) diode. The magnitude per cycle of flybackcurrent from NSME is greater for a given input magnetizing forcerelative to the prior art. (See FIG. 5) For example, Kool Mu torroids(Materials from Magnetics) are suitable for this application. Thismaterial is not identified by way of limitation. The material comprises,by weight, 85% iron, 6% aluminum, and 9% silicon. Further, the magneticelement may be air, (permeablity=1); a molypermalloy powder, (MPP) ahigh flux MPP, a powder, a gapped ferrite, a tape wound, a cut magneticelement, a laminated, or an amorphous magnetic element. Unlike the priorart, the NSME is temperature tolerant in that the critical parameters ofpermeability and saturability remain substantially unaffected duringextreme thermal operation over time. Some materials such as air alsoexhibit little or no change in permeability or saturation levels overtime, temperature, and conditions. The prior art uses high permeabilitysaturable materials often greater than 2000 u permeability. Thesemagnetics exhibit undesirable changes in permeability and saturationduring operation at or near rated output making operation at high powerlevels and temperature difficult. See the permeability vs. temperatureFIG. 11. This deficiency is overcome by the use of expensive oversizedmagnetic elements or output current sharing with multiple supplies. (Seethe graph b_(sat) vs. temperature FIG. 12) This invention takesadvantage of the desirable properties of NSME. See the permeability vs.temperature FIG. 17. Prior art saturating magnetic element commonly isoperating at frequencies greater than 500 KHz to achieve greater powerlevels. As a result practitioners experience exponentially greater corelosses (see FIG. 12A) at high frequencies. NSME support operation atlower frequencies 20-600 KHz further reducing switching losses andmagnetic element losses allowing operation at even higher temperatures.See the loss density vs. flux density FIG. 16. Unlike the prior art, theinstant invention uses voltage mode control with over-current shutdown.Material selection is also based upon mass and efficiency. By increasingthe mass of the magnetic element, more energy is coupled moreefficiently. Since there are reduced losses, the dissipation profilefollows I2R/copper losses. The magnetic element is operated at dutycycles of 0%+ to 90%, which, when used to control the primary sidepush-pull voltage, results in efficiencies on the order of 90%.

[0118]FIG. 18A is a schematic representation of the NSME PFT1ASub-circuit transformer PFT1A consists of a primary winding 100 around aNSME 101 with a center-tapped winding 102. TABLE FIG. 18A ElementValue/part number 100 55 turns 230 uh 101 2 × 77932-A7 102 14 turns

[0119] The primary winding 100 has nodes P1B and P1A for connections toexternal AC source. Secondary 102 winding has center-tapped node S1CTand node S1H and S1L connections to the upper and lower halvesrespectively. Winding 102 are typically connected to external full-waverectifier assemblies. Magnetic element 101 comprises a non-saturating,low permeability magnetic material. The permeability is on the order of26 u with a range of 1 u to 550 u, as compared to the prior art, whichis on the order of 2500 u.

[0120] Flyback management is of concern when using such a magneticelement because the magnetic element generates high drain sourcevoltages across the primary switch during the reverse recovery time ofthe flyback diode. Flyback current is available for longer periods afterthe primary switch opens. (See FIG. 5) For example, Kool Mu (Materialsfrom Magnetics are suitable for this application. This material is notidentified by way of limitation. The material comprises, by weight: 85%iron, 6% aluminum, and 9% silicon. Further, the magnetic element may beair; (air magnetic element permeablity=1); a molypermalloy powder (MPP)magnetic element; a high flux MPP magnetic element; a powder magneticelement; a gapped ferrite magnetic element; a tape wound magneticelement; a cut magnetic element; a laminated magnetic element; or anamorphous magnetic element. During operation the temperature of the NSMErises, the permeability slowly decreases, thereby increasing thesaturation point. Some materials such as air exhibit no or very smallchanges in permeability or saturation levels. Unlike prior art usinghigh permeability materials greater than 2000 u permeability rapidlyincreases at high temperatures. See the permeability vs. temperatureFIG. 11. Prior art also suffers from reduced magnetic element saturationlevels at high temperatures, making operation at high power levels andtemperature difficult and may require the use of expensive oversizedmagnetic elements. See the graph b_(sat) vs. temperature FIG. 12 thisinvention takes advantage of the desirable NSME properties. See thepermeability vs. temperature FIG. 17. Operation at lower frequencies20-600 KHz reduces switching losses and magnetic element losses allowingoperation at higher temperatures. See the loss density vs. flux densityFIG. 16. Unlike the prior art, the instant invention uses voltage modecontrol with over-current shutdown. Material selection is also basedupon mass and efficiency. By increasing the mass of the magneticelement, more energy is coupled more efficiently. Since there arereduced losses, the dissipation profile follows I2R/copper losses. Themagnetic element is operated at duty cycles of 0%+ to 90%, which, whenused to control the primary side push-pull voltage, results inefficiencies on the order of 90%. FIG. 18B is a schematic representationof the NSME BL1, Sub-circuit BL1, consists of a winding 100 around aNSME 101. TABLE FIG. 18 Element Value/part number 107 40 turns 50 uh 1012 × 77932-A7

[0121] Magnetic element BL1, may also be constructed from one or moremagnetic elements in series or parallel. Assuming minimal mutualcoupling the total inductance is the arithmetic sum of the individualinductances. For elements in parallel the (assuming minimal mutualcoupling) the total inductance is the reciprocal of the arithmetical sumof the reciprocal of the individual inductances. In this way multiplemagnetic elements may be arranged to meet packaging, manufacturing, andpower requirements. The primary winding 100 has nodes P2B and P2A forconnections to external AC source. Magnetic element 101 comprises anon-saturating, low permeability magnetic material. The permeability ison the order of 26 u with a range of 1 u to 550 u, as compared to theprior art, which is on the order of 2500 to 5000 u. Flyback managementis of concern when using such a magnetic element because the magneticelement generates high drain source voltages across the primary switchduring the reverse recovery time of the flyback diode. Flyback currentis available for longer periods after the primary switch opens. (SeeFIG. 5) For example, Kool Mu (Materials from Magnetics are suitable forthis application. This material is not identified by way of limitation.The material comprises, by weight: 85% iron, 6% aluminum, and 9%silicon. Further, the magnetic element may be air (air magnetic elementpermeablity=1); a molypermalloy powder (MPP) magnetic element; a highflux MPP magnetic element; a powder magnetic element; a gapped ferritemagnetic element; a tape wound magnetic element; a cut magnetic element;a laminated magnetic element; or an amorphous magnetic element. Duringoperation temperature of the magnetic element rises, the permeabilityslowly decreases, thereby increasing the saturation point. Somematerials such as air exhibit no or very small changes in permeabilityor saturation levels. Unlike prior art using high permeability materialsgreater than 2000 u permeability rapidly increases at high temperatures.See the permeability vs. temperature FIG. 11. Prior art also suffersfrom reduced magnetic element saturation levels at high temperatures,making operation at high power levels and temperature difficult and mayrequire the use of expensive oversized magnetic elements. (See the graphb_(sat) vs. temperature FIG. 12) This invention takes advantage of thedesirable NSME properties. (See the permeability vs. temperature FIG.17.) Prior art often operates at high switching frequencies 100-1000 kHzto avoid the saturation problem. Only to increase switching and corelosses. (See FIG. 12A) This inventions use of the desirable NSMEproperties allows operation at lower frequencies 20-600 KHz furtherreducing switching losses and magnetic element. See the loss density vs.flux density FIG. 16. Unlike the prior art, the instant invention usesvoltage mode control with over-current shutdown. Material selection isalso based upon mass and efficiency. By increasing the mass of themagnetic element, more energy is coupled more efficiently. Since thereare reduced losses, the dissipation profile follows I2R/copper losses.

[0122]FIG. 18C is a schematic representation of a distributed NSMEPFT1D. This is shown to exemplify distributed magnetics enableadvantageous high voltage converter design variations that support formfactor flexibility and multiple parallel secondary outputs from seriescoupled voltage divided primary windings across multiple NSME. Thismagnetic strategy is useful in addressing wire insulation, form factorand packaging limitations, circuit complexity and manufacturability. Inthis example a 500W converter is required to fit in a low profilepackage. Sub-circuit PFTD1 consists of three magnetic elements 120, 121and 124 with series connected primaries. TABLE FIG. 18C ElementValue/part number 113 77352-A7 122 23 Turns 123 23 Turns 112 67 uh (55turns) 114 77352-A7 116 67 uh (55 turns) 117 77352-A7 118 67 uh (55turns)

[0123] AC voltage is applied to 112 pin P1B then from P1C throughconductor 115 to 116 pin P1D. Winding 116 pin P1E is connected throughconductor 119 to 118 pins P1F then to pin P1A. Original Sub-circuit PFT1(FIG. 18) consists of a primary winding 100 around a NSME 101 with twocenter-tapped windings 122 and 123. By way of example sub-circuit PFT1Dwill be implemented as three magnetic elements. For a 500-wattexpression a total inductance of 203 uH is required in winding 100 (FIG.18). Dividing the primary inductance by the number of elements, in thiscase three requires elements 112, 116 and 118 have 67 uH of inductance.The energy storage is equally distributed over the magnetic assembly120, 121 and 124. The 500 watt converter in (FIG. 1) employs two (KoolMu part number 77932-A7) 0.9 oz (25 gram) NSME to form 101 (FIG. 18).Sub-circuit PFT1 magnetic element 101 (FIG. 18) may be expressed asthree 0.5-0.7 oz (14-19 gram) elements. Three 0.5-oz Kool Mu elements(part number 77352-A7) were selected. To realize 67 uH of primaryinductance 55 turns are required for elements 112, 116 and 118. Theprimary circuit has nodes P1B and P1A for connections to external ACsource. Secondary 102 winding has center-tapped node S1CT and node S1Hand S1L connections to the upper and lower halves respectively.Secondary 123 winding has center-tapped node S2CT and node S2H and S2Lconnections to the upper and lower halves respectively. Both 122 and 123are connected to external full-wave rectifier assemblies. Magneticelement magnetic element 120, 121 and 124 comprises a non-saturating,low permeability magnetic material. The permeability is on the order of26 u with a range of 1 u to 550 u, as compared to the prior art, whichis on the order of 2500 u. Flyback management is of concern when usingsuch a magnetic element because the magnetic element generates highdrain source voltages across the primary switch during the reverserecovery time of the flyback diode. Flyback current is available forlonger periods after the primary switch opens. (See FIG. 5) For example,Kool Mu (Materials from Magnetics are suitable for this application.This material is not identified by way of limitation. The materialcomprises, by weight: 85% iron, 6% aluminum, and 9% silicon. Further,the magnetic element may be air (air magnetic element permeablity=1); amolypermalloy powder (MPP) magnetic element; a high flux MPP magneticelement; a powder magnetic element; a gapped ferrite magnetic element; atape wound magnetic element; a cut magnetic element; a laminatedmagnetic element; or an amorphous magnetic element. During operation thetemperature of the NSME, the permeability slowly decreases, therebyincreasing the saturation point. Some materials such as air exhibit noor very small changes in permeability or saturation levels. Unlike priorart using high permeability materials greater than 2000 u permeabilityrapidly increases at high temperatures. See the permeability vs.temperature FIG. 11. Prior art also suffers from reduced magneticelement saturation levels at high temperatures, making operation at highpower levels and temperature difficult and may require the use ofexpensive oversized magnetic elements. (See the graph b t vs.temperature FIG. 12) This invention takes advantage of the desirableNSME properties. See the permeability vs. temperature FIG. 17. Prior artsaturating magnetic element commonly is operating at frequencies greaterthan 500 KHz to achieve greater power levels. As a result practitionersexperience exponentially greater core losses (see FIG. 12A) at highfrequencies. NSME allows operation at lower frequencies 20-600 KHzfurther reduces switching losses and magnetic element losses allowingoperation at even higher temperatures. (See the loss density vs. fluxdensity FIG. 16) Unlike the prior art, the instant invention usesvoltage mode control with over-current shutdown. Material selection isalso based upon mass and efficiency. By increasing the mass of themagnetic element, more energy is coupled more efficiently. Since thereare reduced losses, the dissipation profile follows I2R/copper losses.The magnetic element is operated at duty cycles of 0%+to 90%, which,when used to control the primary side push-pull voltage, results inefficiencies on the order of 90%.

[0124]FIG. 19 is a schematic representation of the non-saturatingpush-pull magnetic element sub-circuit PPT1

[0125] Sub-circuit PPT1 consists of a center-tapped primary winding 104around a NSME 106 with one secondary center-tapped winding 105. TABLEFIG. 19 Element Value/part number 106 77259-A7 105 10 Turns 104 70 Turns

[0126] The primary winding 104 has nodes P2H and P2L for connections toexternal AC sources, and common center-tap node P2CT. Secondary 105winding has center-tapped node SCT and nodes SH and SL connections tothe upper and lower halves respectively. The invention is not limited toa single output. More secondary windings may be added for additionaloutputs. Secondary 105 is connected to an external full-wave rectifierassembly (Example FIG. 25 or 26). The magnetic element magnetic element106 comprises a non-saturating, low permeability magnetic material. Thepermeability is on the order of 26 u with a range of 1 u to 550 u, ascompared to the prior art, which is on the order of 2500 u. Flybackmanagement is of concern when using such a magnetic element as highdrain source voltages across the primary switch are generated during thereverse recovery of the flyback diode. The falling flyback current isavailable for a longer period. (See FIG. 5) For example, Kool Mu(magnetic elements from Magnetics are suitable for this application.This material is not identified by way of limitation. The materialcomprises, by weight; 85% iron, 6% aluminum, and 9% silicon. Further,the magnetic element may be air (comprise an air magnetic element); amolypermalloy powder (MPP) magnetic element; a high flux MPP magneticelement; a powder magnetic element; a gapped ferrite magnetic element; atape wound magnetic element; a cut magnetic element; a laminatedmagnetic element; or an amorphous magnetic element. During operation thetemperature of the NSME rises, the permeability slowly decreases,thereby increasing the saturation point. Unlike prior art using highpermeability materials greater than 2000 u permeability rapidlyincreases at high temperatures. See the permeability vs. temperatureFIG. 11. Prior art also suffers from reduced magnetic element saturationlevels at high temperatures, making operation at high power levels andtemperature difficult and may require the use of expensive oversizedmagnetic elements. (See the bsat vs. temperature FIG. 12) This inventiontakes advantage of the desirable NSME properties. (See the permeabilityvs. temperature FIG. 17) Operation at lower frequencies 20-600 KHzreduces switching losses and magnetic element losses allowing operationat higher temperatures. See the loss density vs. flux density FIG. 16.Unlike the prior art, the instant invention uses voltage mode controlwith over-current shutdown. Material selection is also based upon massand efficiency. By increasing the mass of the magnetic element, moreenergy is coupled more efficiently. Since there are reduced losses, thedissipation profile follows I2R/copper losses. The magnetic elementprimary is driven in a push-pull fashion at a duty cycle of 48-49%resulting in efficient use of the magnetic element volume.

[0127]FIG. 19A is a schematic representation of the non-saturatingpush-pull magnetic element sub-circuit PPT1. Sub-circuit PPT1 consistsof a center-tapped primary winding 134 around a NSME 136 with onesecondary center-tapped winding 135. TABLE FIG. 19A Element Value/partnumber 136 77259-A7 135 10 Turns 134 70 Turns

[0128] The primary winding 134 has nodes P2H and P2L for connections toexternal AC sources, and common center-tap node P2CT. Secondary 135winding has center-tapped node SCT and nodes SH and SL connections tothe upper and lower halves respectively. The invention is not limited toa single output winding. More secondary windings may be added foradditional outputs. Secondary 135 is connected to an external full-waverectifier assembly such as OUTA (FIG. 25), OUTB (FIG. 25A) and OUTBB(FIG. 25B). The magnetic element 136 comprises a non-saturating, lowpermeability magnetic material. The permeability is on the order of 26 uwith a range of 1 u to 550 u, as compared to the prior art, which is onthe order of 2500 u. Flyback management is of concern when using such amagnetic element as high drain source voltages across the primary switchare generated during the reverse recovery of the flyback diode. Thefalling flyback current is available for a longer period. (See FIG. 5)For example, Kool Mu (magnetic elements from Magnetics are suitable forthis application. This material is not identified by way of limitation.The material comprises, by weight; 85% iron, 6% aluminum, and 9%silicon. Further, the magnetic element may be air (comprise an airmagnetic element); a molypermalloy powder (MPP) magnetic element; a highflux MPP magnetic element; a powder magnetic element; a gapped ferritemagnetic element; a tape wound magnetic element; a cut magnetic element;a laminated magnetic element; or an amorphous magnetic element. Duringoperation the temperature of the NSME rises, the permeability slowlydecreases, thereby increasing the saturation point. Unlike prior artusing high permeability materials greater than 2000 u permeabilityrapidly increases at high temperatures. See the permeability vs.temperature FIG. 11. Prior art also suffers from reduced magneticelement saturation levels at high temperatures, making operation at highpower levels and temperature difficult and may require the use ofexpensive oversized magnetic elements. (See the bsat vs. temperatureFIG. 12) This invention takes advantage of the desirable NSMEproperties. (See the permeability vs. temperature FIG. 17) Operation atlower frequencies 20-600 KHz reduces switching losses and magneticelement losses allowing operation at higher temperatures. See the lossdensity vs. flux density FIG. 16. Unlike the prior art, the instantinvention uses voltage mode control with over-current shutdown. Materialselection is also based upon mass and efficiency. By increasing the massof the magnetic element, more energy is coupled more efficiently. Sincethere are reduced losses, the dissipation profile follows I2R/copperlosses. The magnetic element primary is driven in a push-pull fashion ata duty cycle of 48-49% resulting in efficient use of the magneticelement volume.

[0129]FIG. 20 lightning input protection and filter sub-circuit LL

[0130]FIG. 20 is a schematic showing the inventions filter and lightninginput protection circuit for an AC line connected converter. Theprotection sub-circuit LL comprises a Spark gap A1, diodes D20 and D21,capacitor C1 and magnetic elements L1 and L2. TABLE FIG. 20 ElementValue/part number L1 375 uH L2 375 uH C61 0.01 uF C60 0.01 uF A1 400 VSpark Gap C1 0.1 uF D20 1000 V/25A D21 1000 V/25A D22 1000 V/25A D231000 V/25A C2 1.8 uf

[0131] The AC line is connected to node LL2. The common inputfrequencies of DC to 440Hz may be extended beyond this range withcomponent selection. Node LL2 is connected to NSME L1 then to node LL5,the spark gap A1, anode of diode D22 and the cathode of diode D20.Filter capacitor C60 is connected between node LL0 and LL6. Filtercapacitor C61 is connected between node LL0 and LL5. The low side of ACline is connected to node LL1 then to magnetic element L2 the other sideL2 is connected to spark gap Al, anode of diode D23 and the cathode ofdiode D21 and to node LL6. Capacitor C1 is connected to earth ground C1attenuates noise generated by the converter. The use of non-saturatingmagnetic allows the input magnetic elements to absorb very largevoltages and currents commonly generated by lightning, often withoutcausing spark gap A1 to clamp. During UL testing sixty 16ms 2000V pulseswere applied between LL1 and LL2 without realizing spark gap A1 wasmissing with out damage. During normal operation the NSME L1 fluxdensity is a few hundred gauss. The 75 u material from the graph of FluxDensity v. Magnetizing Force (FIG. 15) will accept flux densities atleast 50 times greater with out limitation. This is an example of themagnetic elements ability to perform well at flux densities many timesgreater than prior art. Elements L1 and L2 will block differential orcommon mode line transients. In the event of a very large or longduration line to neutral transient, spark gap A1 will clamp the voltageto a safe level of about 400V. The NSME L1 and L2 have the added benefitof reducing conducted noise generated by the converter.

[0132]FIG. 21 alternate lightning input protection sub-circuit LLA

[0133]FIG. 21 is a schematic showing the inventions alternate lightningprotection sub-circuit for an AC line connected converter. Theprotection circuit comprises a fuse F1, Spark gap A1, capacitors C1, C60and C61 and NSME L3. TABLE FIG. 21 Element Value/part number L3 750 uHC61 0.01 uF C60 0.01 uF F1 10A A1 400 V Spark Gap C1 0.1 uF D20 1000V/25A D21 1000 V/25A D22 1000 V/25A D23 1000 V/25A C2 1.8 uf

[0134] High side of AC line is connected to node LL2, fuse F1 the loadside of the fuse is connected to NSME L3 and to capacitor C61. The loadside L3 is connected to spark gap A1 and the cathode of diode D20 andanode of D22 forming node LL5. The low side of AC line is connected tonode LL6, capacitor C60, spark gap A1, and the cathode of diode D21 andanode of D23. The anodes of diodes D20 and D21 are connected toCapacitor C1. Capacitor C1 is connected to earth ground. C1 attenuatesradiated noise or EMI generated by the converter. The cathode of diodesD22 and D23 are connected to Capacitor C2. Capacitor C2 decouples highfrequency harmonic currents from the line. Capacitors C1, C61 and C60are connected to earth ground node LL0. The use of non-saturatingmagnetics allows the input magnetic element to absorb very largevoltages and currents commonly generated on the AC line by lightning. Atransient on the AC line will be limited by capacitors C60 and C61 andblocked by the non-saturating magnetic L3. In the event of a very largeor long duration line to neutral transient, magnetic element L3 willallow the voltage to rise across spark gap A1, the spark gap will clampthe voltage to a safe level protecting the rectifier diodes D20-D23. TheNSME L3 has the added benefit of reducing conducted noise generated bythe converter. C1 connected to the ground plane is effective inattenuating conducted and radiated EMI.

[0135]FIG. 22 AC line rectifier sub-circuit BR

[0136]FIG. 22 is a schematic showing the inventions AC line rectifier.The rectifier sub-circuit BR1 comprises diodes D20, D21, D22 and D23 andcapacitor C2. TABLE FIG. 22 Element Value/part number D20 1000 V/25A D211000 V/25A D22 1000 V/25A D23 1000 V/25A C2 1.8 uf

[0137] An AC or DC signal from the input filter is connected to bridgerectifier to nodes BR1 and BR2. Node BR1 connects diode D22 anode to D20cathode. Node BR2 connects diode D23 anode to D21 cathode. Node BR+connects diode D22 cathode to D23 cathode. Node BR− connects diode D20anode to D21 anode. The common input frequencies of DC to 440 Hz may beextended beyond this range with component selection. Capacitor C2 isselected to improve power factor for a particular operating frequencyand to de-couple switching currents from the line. Diodes are selectedto reliably block the expected line voltage and current demands of thenext converter stage.

[0138]FIG. 23 controller sub-circuit PFA

[0139]FIG. 23 is the inventions AC to DC controller sub-circuit.Sub-circuit PFA consists of resistors R313 and R316, capacitors C308,and C313 and PWM controller IC U1A. TABLE FIG. 23 Element Value/partnumber C311 0.1 uf C308 .01 uf R313 15k ohms R316 15k ohms C313 4700 pfR308 25k ohms U1A MIC38C43 (Micrel)

[0140] Control element U1A connects to a circuit with the followingseries connections: from pin 1 to feedback node/pin PF1 then tocapacitor C308 then to the pin 2 node of U1A. Internal 5.1-voltreference U1A pin 8 or node PFA2 through resistor R308 to the pin 4node. U1A pin 4 is connected through capacitor C313 to return node BR−.This arrangement allows the PFC output to be pulse width modulated withapplication of voltage to PF1. External feedback current applied to U1Apin 1 and node PF1. Node PFVC is connected to resistor R313 to pin 3 ofU1A. Resistor R316 is connected to pin 3 then to return node BR−. Powerpin 7 is connected to node PFA+. Control element switch drive U1A pin 6is connected to node PFCLK. Return ground node of U1A pin 5 is connectedto return node BR−. In the event of a component failure in the primaryfeed networks such as IPFFB (FIG. 40), FBA (FIG. 40A), IFB (FIG. 40B)and FB1 (FIG. 41). The output voltage of the boost stage may rapidlyincrease to destructive levels. Fast over voltage feedback networksIOVFB (FIG. 40C) or OVP2 (FIG. 42B) increases the current into PF1thereby limiting the output voltage to a safe level. In additionlatching type over voltage protection networks such as OVP (FIG. 42),OVP1 (FIG. 42A) and OVP2 (FIG. 42B) maybe used. The latching type killspower to the control circuit thereby stopping the boost action. Thelatching type networks require power to be cycled to the converter toreset the latch. IFB Input node PFVC is connected to resistor R313 tointernal zero crossing detector connected to pin 3 and through R316 toreturn node BR−. PFVC is connected to a magnetic element windingreferenced to BR−. A new conduction cycle is started each time the biasin the magnetic element goes to zero. Power factor corrected is realizedby chopping the input at a high frequency. The average pulse widthdecreases at higher line voltage and increases at lower voltage for agiven load. Frequency is lower at line peaks and higher around zerocrossings. In this way the converter operates with a high input powerfactor.

[0141]FIG. 24 Controller with power factor corrected sub-circuit PFBFIG. 24 is the alternate power factor controller sub-circuit.Sub-circuit PFB consists of resistors R313, R339, R314, R315, R328,R340, R341 and R346, diode D308, capacitors C310, C318, C338, C340, C341and C342, transistor Q305, and control element IC U1B. TABLE FIG. 24Element Value/part number Q305 BCX70KCT R339 432k ohms C338 0.22 uf C3180.22 uf R314 2.2 MEG ohms R315 715k ohms C341 0.33 uf C342 0.01 uf C3400.001 uf R328 1 MEG ohms R346 7.15k ohms D308 10BQ040 R340 449k ohmsR313 22k ohms U1B MC34262 (Motorola) R341 499k ohms

[0142] Control element U1B connects to a circuit with the followingseries connections: from pin 1 to node/pin PF1 to capacitor C338 inseries with resistor R339, and then to the pin 2 node of U1B. Pin 1 isthe input to an internal error amplifier and connection to externalfeedback networks. (See FIG. 40, 40A, 40B, 40C. and 41) Increasing thevoltage on pin 1 decreases the pulse width of the PFCLK node pin 7.Resistor R328 is connected to the fullwave rectified AC line haversinevoltage on node BR+ then to U1B pin 3 and then to resistor R346 inparallel with capacitor C342 to return node BR−. Node PFSC connects toseries resistors [R341+R340] which are connected to U1B pin 4 then todiode D308 in parallel with capacitor C340 to return node BR−. Power toPFC controller is applied to node PFB+ and to U1B pin 8. Output clocknode PFCLK is connected to U1B pin 7, to external buffer sub-circuit AMP(FIG. 29). Transistor Q305 collector is connected to the pin 2 node ofU1B. The base is connected in series through resistor R314 to capacitorC318, then to the pin 2 node of U1B. The base is also connected to[C310∥R315], then to return node BR−. Emitter of Q305 is connected toreturn node BR−. Transistor Q305 provides a soft start compensation rampto the controller error amp reducing the stress and DC overshoot in theconverter at power up. Capacitor C341 is connected from U1 pin 2 toreturn node BR−. U1B pin 1 is connected to pin PF1, capacitor C338 inseries with resistor R339 to transistor Q305 collector and to U1 pin 2.Current switched by PFC power switch Q1 (FIG. 4 & 3) is sensed by R26(see FIG. 4). Series resistors [R340+R341] to U1B pin 4 connect voltagedeveloped across R26. This voltage is compared to an internal 1.5-voltreference, comparator output turns off the switch drive pin 7 of U1Bduring times of high current that occur during startup or during veryhigh load or low line conditions. Capacitor C340 is connected between U1pin 4 to return node BR− filter high frequency components. Schottkydiode D308 connected between U1 pin 4 to return node BR− protects thecontroller (U1 pin 4) substrate from negative current injection. Maximumswitch current value is set by R26 over currents are automaticallylimited in each cycle by the PFC controller. The rectified fullwavehaversine at pin 3 of U1B is multiplied by the error voltage on pin 2.The product is compared to the magnetic element/switch current measuredby R26 on pin 4. Gate drive on pin 7 turns off when the sensed magneticelement current increases to the current comparator level. This actionhas the effect of modulating the switch Q1 “on” time tracking the ACline voltage. External feedback networks are connected to node PF1. Inthe event of a component failure in the primary feed network such asIPFFB (FIG. 40), FBA (FIG. 40A), IFB (FIG. 40B) and FB1 (FIG. 41). Theoutput voltage of the boost stage may rapidly increase to destructivelevels. Fast over voltage feedback networks IOVFB (FIG. 40C) or OVP2(FIG. 42B) increases the current into PF1 thereby limiting the outputvoltage to a safe level. In addition latching type over voltageprotection networks such as OVP (FIG. 42), OVP1 (FIG. 42A) and OVP2(FIG. 42B) maybe used. The latching type removes power to the controlcircuit thereby stopping the boost action. The latching type networksrequire power to be cycled to the converter to reset the latch.Modulating the voltage at PF1 changes the duty cycle of the PFC and thefinal output voltage. In this way the PFC may be used as a pre-regulatorto additional output stages.

[0143]FIG. 25 Output rectifier and filter sub-circuit OUTA

[0144]FIG. 25 is a schematic of a full wave rectified output stage andfilter sub-circuit OUTA. The rectifier stage consists of diodes D80 andD90. The filter consists of resistor R21, magnetic element L30 andcapacitors C26, C27, C28, C29, C30, C31 and C32. Table ElementValue/part number D80 40CTQ150 D90 40CTQ150 R21 100 ohms C26 500 pf C27200 pf C28 0.1 uf C29 10,000 uf C30 10,000 uf C31 0.1 uf C32 200 pf L3010 uh

[0145] Input node/pin C7B is connected to the high side of externalcenter-tapped magnetic element secondary winding.

[0146] Node C7B connects to anode of diode D8 and to capacitors C26 andC27 in the following arrangement. Capacitor C27 is connected acrossdiode D80, capacitor C26 is connected in series to R21. Input node/pinC8B is connected to the low side of external center-tapped magneticelement secondary winding. Pin C8B is connected to anode of diode D9 andto resistor R21, capacitor C32 is connected across diode D90. CapacitorsC27 and C32 is a small value to reduce high frequency noise generated byrapid switching the high speed rectifier D80 and D90 respectively.Capacitor C26 and resistor R21 are used to further dissipate highfrequency energy. Anode of diodes D80 and D90 is connected to parallelcapacitors C28∥C29 and NSME L30. Capacitors C28 and C31 are soliddielectric types selected for low impedance to high frequency signals.Capacitors C29 and C30 are larger polarized types selected for lowimpedance at low frequencies and for energy storage. Magnetic element L3is connected to diode D8 cathode the second terminal of L30 is connectedto parallel capacitors C31 and C30 and pin OUT+. Node OUT+ is the outputpositive and is connected to external feedback sense line to isolatedfeedback network. The other side of parallel capacitors[C28∥C29∥C30∥C31] is connected to pin OUT− and the center-tap of themagnetic element secondary forming the return node. The combination ofcapacitors [C2881 C29], L30 and capacitors [C30∥C31] form a low pass pitype filter. Sub-circuit OUTA performs efficient fullwave rectificationand filtering.

[0147]FIG. 25A Output rectifier sub-circuit OUTB

[0148]FIG. 25A is a schematic of a full wave rectified output stage. Therectifier stage consists of diodes D80 and D90 and capacitors C931 andCG928. Table Element Value/part number D80 40CTQ150 D90 40CTQ150 C928.01 uf C931 10,000 uf

[0149] Input node/pin C7B is connected to high side of externalcenter-tapped magnetic element secondary winding. Node C7B connects toanode of diode D80. Input node/pin C8B is connected to low side ofexternal center-tapped magnetic element secondary winding is connectedto anode of diode D90. Node OUT− is the negative output and return lineto the external isolated feedback network and load not shown. Cathodesof diodes D80 and D90 are connected to parallel capacitors C931 andC928. Capacitor C928 is a solid dielectric type selected for lowimpedance to high frequency signals. Capacitor C931 is a largerpolarized selected for low impedance to low frequency signals and forenergy storage. Node OUT+ is the output positive and is connected toexternal feedback sense line to isolated feedback network. The otherside of parallel capacitors C928∥C931 is connected to the center-tap ofthe magnetic element secondary forming the node OUT−. The use of theNSME for the push-pull magnetic element requires only minimal filteringafter the rectifiers.

[0150]FIG. 25B AC rectifier and filter sub-circuit OUTB

[0151]FIG. 25B is a schematic diagram of an alternate final outputrectifier and filter sub-circuit OUTB. The rectifier sub-circuit OUTBcomprises diodes D40, D41, D42 and D43 and capacitor C931 and C928. FIG25B Table Element Value/part number D40 40CTQ150 D41 40CTQ150 D4240CTQ150 D43 40CTQ150 C928 .01uf C931 10,000 uf

[0152] An AC or DC signal is connected to nodes C7B and C8 b. Node C7Bconnects diode D41 anode to D40 cathode. Node C8b connects diode D42anode to D43 cathode. Node OUT+ connects diode D42 cathode to D43cathode. Node OUT− connects diode D40 anode to D43 anode. Diodes areselected to reliably block the expected line voltage and current demandsof the load. For low voltage outputs, Schottky type diodes are used dueto their low forward voltage drop. Higher voltages would use high-speedsilicon diodes due to their ability to withstand high peak inversevoltage (PIV). The use of the NSME for the push-pull magnetic elementrequires only minimal filtering after the rectifiers. Capacitor C928 isshown schematically as a single device. Capacitor C931 is a largerpolarized selected for low impedance to low frequency signals and forenergy storage a typical value may be 200 uF. To increase thecapacitance or reduces the output impedance multiple capacitors may beused. C931 is a solid dielectric type and is selected for it's lowimpedance to high frequencies. As is selected to reduces noise for aparticular operating frequency and power level. Capacitor C928 isselected for the operating frequency and power level. Sub-circuit OUTBperforms AC to DC rectification and filtering at slightly lowerefficiency due to the extra junctions.

[0153]FIG. 26 Floating 18_Volt DC control power sub-circuit CP

[0154] Sub-circuit CP consists of diodes D501, D502 and D503, resistorR507, regulator Q504, and capacitors C503, C504, C505, C506, and C507.Table Element Value/part number C503 .33 uF C504 100 uF D501 MURS120T3C505 .33 uf Q504 LM7818A C508 100 uf C507 100 uf D503 MURS130T3 D502MURS120T3

[0155] Node CT1A connects to anode of D503 and to the upper externalcenter tapped secondary winding. Node CT2A connects to anode of D502 andto the lower external center tapped secondary winding. Node CT0 connectsto the external winding center tap. Node CT0 is also the return line andit connects to Q504 pin 2, and capacitors C503, C504, C505, C506, andC507. The cathode of each of diodes D502 and D503 is connected toresistor R507. R507 is then connected to the pin 1 (input) node ofvoltage regulator Q504. Voltage regulator Q504 Pin 3 is the 18 vdcregulated DC output is connected to the anode of blocking diode D501.Three-pin voltage regulator Q504 is of the type LM7818 a common devicemade by a number of manufacturers. Capacitors C503, C505, C506 are 0.1uF solid dielectric type and are used to filter high frequency rippleand to prevent Q504 from oscillating. The junction of C503, C504 andD501 cathode is the output node CP1+. Isolated 18-volt DC is availablebetween nodes CT0 and CP+. Used for regulator circuits and output switchdrive during normal operation.

[0156]FIG. 27 second Floating 18_-Volt DC push-pull control powersub-circuit CPA. Sub-circuit CPA consists of diodes D601, D602 and D603,resistor R607, regulator B604 and capacitors C603, C604, C605, C606,C607 and C608. Table Element Value/part number C603 .33 uF C604 100 uFD601 MURS120T3 C605 .33 uF Q604 LM7818A C608 100 uF C607 .22 uF R607 7.5ohms D603 MURS120T3 D602 MURS120T3

[0157] Node CT1B connects to anode of D603 and to the upper externalcenter tapped secondary winding. Node CT1B connects to anode of D602 andto the lower external center tapped secondary winding. Node CT20connects to the external winding center tap. Node CT0 is also the returnline and it connects to Q604 pin 2, and capacitors C603, C604, C605,C606, and C607. The cathode of each of diodes D602 and D603 is connectedto resistor R607. R607 is then connected to the pin 1 (input) node ofvoltage regulator Q604. Voltage regulator Q604 Pin 3 is the 18 vdcregulated DC output and is connected to the anode of blocking diodeD601. Capacitors C603, C605, C606 are solid dielectric type and are usedto filter high frequency ripple and to prevent Q604 from oscillating.The junction of C603, C604 and D601 cathode is the output node CP1+.Isolated 18-volt DC is available between nodes CT20 and CP2+. To be usedfor regulator circuits and output switch drive during normal operation.

[0158]FIG. 28 over temperature protection sub-circuit OTP

[0159]FIG. 28 is the main switch over temperature protection sub-circuitOTP. The sub-circuit OTP comprises thermal switch and resistors R711 andR712. Table Element Value/part number THS1 67F105 (105C) R711 20 ohmsR712 20 ohms

[0160] Gate drive power is applied to input node GAP and to thermalswitch THS1. Maximum FET gate voltage requires the input power voltagebe less than 20 volts, the voltage selected was 18 volts. The other sideof THS1 is connected to parallel resistors [R711∥R712]. A singleresistor may represent the resistors. The figure depicts the surfacemount arrangement. The other side of [R711∥R712] connects to output nodeTS+. Normally closed thermal switch TS1 is in contact with main switchtransistor Q1. In the event of temperature greater than 105 C. THS1opens, thus removing power to the buffer sub-circuit AMP1 (FIG. 29)causing switch Q1 to default to a blocking state protecting the boostswitch should the optional cooling fan fail or the circuit reach hightemperatures. In this instant invention the speed up buffer AMP (FIG.29) non-saturating magnetics (FIG. 18, 18A and 19) allows the mainswitch and to run cooler than prior art for a given power level. Whenswitch temperature returns to normal range THS1 will close, allowing thePFC to resume normal operation. Under normal load and ambienttemperatures the thermal switch THS1 should never open.

[0161]FIG. 29 PFC Buffer Circuit sub-circuit AMP, AMP1, AMP2, AMP3

[0162] Switch drive command from PFCLK (FIG. 23 and 24) or PWFM (FIG.33) control elements are connected to a gate buffer circuit. Thesub-circuit AMP is comprised of power FET Q702, Darlington pair Q703,capacitors C709 and C715, and resistors R710 and R725. Table ElementValue/part number C715 1000 pf C709 33 uF Q702 NOS355NCT Q703 FZT705CTR710 0 ohms R725 22.1k ohms

[0163] DC Power is applied to node GAT+to transistor Q702 drain and tocapacitor C709, which goes to ground. Maximum gate voltage requires theinput power voltage must be less than 20 volts, 18-volts was selected.Input node GA1 is connected to the gate of FET Q702 is connected to thebase of BJT1 of the Darlington pair Q703 and to capacitor C715. C715 isconnected across the Darlington pair from the base, pin 1, to thecollectors, pins 2 and 4, Q703 collector node is also connected toground. The emitter of BJT2 is connected to the gate of FET Q1. Thesource of FET Q702 is connected through small optional series resistorR710 to the gate of the output switch or node GA2. Some power FET'sunder certain load may tend to oscillate when driven from a lowimpedance source such as this buffer. A small resistance ofapproximately 2 ohms or less may be required with out significantslowing of the switch. In most cases R710 is replaced with a zero ohmjumper. Resistor R725 is connected from node GA0 and source of Q702. Theinput switching signal to node GAP is in range of 20 kHz to 600 kHz.Very fast “on” times are realized by proving a low impedance to rapidlycharge the output switch gate connected to node GA2. Capacitor C709provides additional current when Q702 switches on. Transistor Q703provides low impedance to rapidly remove the charge from the gategreatly reducing the “off” time. This particular topology providesoutput switch rise times on the order of 10 ns, as compared to theindustry standard rise time of 250 ns. The corresponding fall time is<10 nS, again as compared to an industry fall time of 200-300 ns (SeeFIG. 13 and 14). In the event the converter is operated at very highambient temperatures a thermal switch may be placed in series with inputpower pin GA+. This allows the switch transistor to be gracefullydisabled. Sub-circuit AMP greatly reduces switching losses allowingconverter operation in some cases with out the common prior art forcedair-cooling.

[0164]FIG. 30 snubber sub-circuit SN

[0165]FIG. 30 is a schematic diagram of a snubber sub-circuit of theinvention. The snubber sub-circuit SN is comprised of diodes D804 andD805 and resistors R800, R817, R818, and capacitors C814 and C819. TableElement Value/part number R800 12 ohms R817 1 mohm R818 1 mohm C814 33pF C819 560 pF D805 MUR160

[0166] Node SNL2 connects to the drain terminal of the external outputswitch and to flyback side of the inductive load. Input node SNL2connects to R800 in series with capacitor C819 to node SNOUT. Diode D805anode is connected to node SNL2 with resistors [R817∥R818] in parallelwith D805. Resistors R817 and R818 may be combined to a single resistor.The cathode of D805 is connected to capacitor C814 that connects tonode/pin SNL1. Node SNL1 connects to the supply side of external loadmagnetic element. The other leg of external magnetic element isconnected to the anode of D805 and the anode side of external flybackdiode D4. The one MEG ohm resistors R817 and R818 bleed the charge fromC814 resetting it for the next cycle. Capacitor C819 and resistor R800captures the high frequency event from the transition of externalflyback diode D4 and moves part of the energy into the external holdupcapacitor connected to node SNOUT. Since external flyback diode D4 andD805 isolate the drain of the output switch, faster switching occursbecause the output switch does not have to slew the extra capacitance ofa typical drain/source connected snubber circuit. Note that this circuitdoes not attempt to absorb the flyback in large RC networks that convertuseful energy to losses. Nor does it attempt to stuff the flyback toground, adding capacitance and slowing the output switch and increasingswitching losses. This sub-circuit is used with it's mirror SNB (FIG.32) across the external push-pull switches. This design returns the someof the flyback energy back to the input supply or output load. The“snubbering” action slows the rise of the flyback giving time for theexternal flyback diode to start conduction. The circuit efficientlymanages high frequency flyback pulses.

[0167]FIG. 30A diode snubber sub-circuit DSN

[0168]FIG. 30A is a schematic diagram of a diode snubber sub-circuit ofthe invention. The snubber sub-circuit DSN is comprised of diodes D51,D52, D53, D54 and D55 and capacitors C51, C52, C53, C54 and C55. TableElement Value/part number C51 220 pf 100 v C52 220 pf 100 v C53 220 pf100 v C54 220 pf 100 v C55 220 pf 100 v D51 Schottky 1-2ns 100 vSMBSR1010MSCT D52 Schottky 1-2ns 100 v SMBSR1010MSCT D53 Schottky 1-2ns100 v SMBSR1010MSCT D54 Schottky 1-2ns 100 v SMBSR1010MSCT D55 Schottky1-2ns 100 v SMBSR1010MSCT

[0169] Pin SNL2 is connected to the anode of D51 the cathode of D51 isconnected to the anode of D52 the cathode of D52 is connected to theanode of D53 the cathode of D53 is connected to the anode of D54 thecathode of D54 is connected to the anode of D55 the cathode of D55 isconnected to pin SNOUT. Capacitors are connected across each diodeforming a series parallel combination of[D51∥C51]+[D52∥C52]+[D53∥C53]+[D54∥C54]+[D55∥C55]. Node SNL2 connects tothe drain terminal of the external output switch and to flyback side ofthe inductive load. The external fly-back rectifier diode D4 (FIG. 1, 3and 4) anode is connected to node SNL2. Node SNOUT connects to thestorage capacitors [C16∥C17] (FIG. 1, 3 and 4) and to the cathode of theflyback diode D4. External diode D4 in parallel with DSN forms a hybriddiode. The Schottky diode has the desirable characteristics of fastrecovery time (less than 6 nanoseconds (6*10 ^ −9)) and low forwardvoltage drop (0.4-0.9 Volts) at high currents. The Schottky diodesuffers from limited reverse blocking voltage currently 100 V maximum.Each diode will block 100V; the parallel capacitors distribute thereverse voltage equally across the diode string. As the reverse junctioncapacitance of each diode is less than 10 pf much smaller than theparallel capacitor. Thus the reverse voltage is nearly equally dividedacross the diodes. To guarantee even voltage division 5% or bettercapacitor matching is required. High precision is common and inexpensivefor small capacitors. Different blocking voltages may be achieved byadjusting the number of diode/capacitor pairs. By way of example not asa limitation 500V was selected. The main fly-back rectifier diode D4will block high voltages but suffers from long reverse recovery time50-500 nanoseconds is common in fast recovery diodes. What is needed isa diode with low voltage drop, high blocking voltage and very shortrecovery time. The snubber DSN in parallel with the main fly-backrectifier comes very close to that ideal diode. The total blockingvoltage is achieved by the adding the individual diode blockingvoltages. The recovery time is determined by the slowest diode in thestring often less than 5 nanoseconds. The low forward voltage drop isachieved when the slower main rectifier begins conduction. Lowcapacitance is also realized, as the capacitance is ⅕ of the individualcapacitors. This hybrid diode begins rectification immediately after themain switch stops conduction and the non-saturating magnetic beginsreleasing its energy. This effectively limits the high voltage flybackover shoot to less than 40-70 volts. This keeps the switch well insideit's safe operating area (SOA) allowing the switch to be run at highervoltages for higher output power and additional efficiency gain, or touse a less expensive lower voltage switch while keeping the same voltagemargins. Since external flyback diode D4 and D805 isolate the drain ofoutput switch, faster switching occurs because the output switch doesnot have to slew the extra capacitance of the typical snubber circuit.Note that this circuit does not attempt to absorb the flyback in largeRC networks that generate additional heat. Nor does it attempt to stuffthe flyback to ground, adding capacitance and slowing the output switch,increasing switching losses. Sub-circuit DSN may be used in parallelwith any slower rectifier such as flyback diode D4 to assist the mainrectifier. This providing additional protection to the switch andrectifying the portion of the flyback pulse before the main rectifierbegins condition. That high frequency energy ends up as heat or radiatednoise.

[0170]FIG. 31 snubber sub-circuit SNA

[0171]FIG. 31 is a schematic diagram of a snubber sub-circuit of theinvention. The snubber sub-circuit SNA is comprised of resistor R810 andR811 and capacitors C820 and C821. Table Element Value/part number R810500 pF C811 330 pF C820 12 ohm C821 10 ohm

[0172] Node SNAL connects to series resistor R810 to capacitor C820 tonode SNA2 then to capacitor C821 and series resistor R811 to node SNA3.Node SNAL connects to the external magnetic element center tap. NodeSNA2 connects to the drain terminal of the external output switch and toflyback side of the inductive load. Node SNA3 connects to the sourceterminal of the external output switch. Resistor R810 and C820 attemptto absorb part of the flyback to reduce voltage transients across theswitch. Part of the flyback is returned to ground by C821. Thissub-circuit is used with it is mirror SNA (FIG. 31) across the externalpush-pull switches. The “snubbering” action slows the rise of theflyback giving time for the external rectifier diodes D8 and D9 of FIG.25 or 25A to start conduction. The circuit efficiently manages highfrequency flyback pulses.

[0173]FIG. 32 snubber sub-circuit SNB

[0174]FIG. 32 is a schematic diagram of a snubber sub-circuit of theinvention. The snubber sub-circuit SNB comprises resistor R820 and R821and capacitors C840 and C841. TABLE FIG. 32 Element Value/part numberC840 500 pF C841 330 pF R820 12 ohm R821 10 ohm

[0175] Node SNB1 connects to series resistor R820 to capacitor C820 tonode SNA2 to capacitor C841 and to series resistor R821 to node SNB3.Node SNB1 connects to the external magnetic element center tap. NodeSNB2 connects to the drain terminal of the external output switch and toflyback side of the inductive load. Node SNB3 connects to the sourceterminal of the external output switch. Resistor R820 and C840 attemptto absorb part of the high frequency flyback to reduce voltagetransients across the switch. C841 and R821 return part of the flybackto ground. The “snubbering” action slows the rise of the flyback givingtime for the external rectifier diodes D8 and D9 of FIG. 25 or 25A tostart conduction. The circuit efficiently manages high frequency flybackpulses.

[0176]FIG. 33 Pulse/Frequency modulator PWFM

[0177]FIG. 33 is the inventions PWM (pulse width modulator) and FM(frequency modulator) sub-circuit. Sub-circuit PWFM consists ofresistors R401, R402, R403, and R404 capacitors C401, C402, C403, C404,C405 and C406, controller IC U400 and diode D401. TABLE FIG. 33 ElementValue/part number R404 50k ohms C406 100 uf C401 0.22 uF C403 0.01 uFC405 2200 pF C404 470 pF C402 0.22 uF R403 50k ohms D401 RLS139 (lowleakage) R401 2.2 MEG ohms R402 150k ohms U400 MIC38C43

[0178] Control element U400 connects to a circuit with the followingseries connections: from pin 1 to feedback pin PW1 then to the wiper ofadjustable resistor R404 to return node PWFM0. Resistor R404 may bereplaced with two fixed resistors. Capacitor C403 is connected from pin2 to pin 1. Capacitor C403 is used to filter the error amp output. Theupper half of resistor R404 is connected to node REF1 pin 8 the 5.0-Voltinternal reference. Internal 5.0-volt reference U400 pin 8 or Node REF1is connected to the upper half of resistor R403 and through capacitorC402 to return node PWFM0. The reference provides current to externalfeed back networks. Wiper of R403 connects to node FM1 to pin 4, throughR402 to pin 3, and through C404 to return node PWFM0. Resistor R403 maybe replaced with two fixed resistors. Pulse width timing capacitor C404connects pin 3 to return node PWFM0. Low leakage diode D401 anode isconnected to pin 3 cathode to output pin 6 node CLK. Resistor R404 setsthe nominal pulse width of output pin 6 node CLK. The pulse width can beadjusted from 0 (off) to 95%. Resistor R403 and C404 determine thenominal operating frequency. With application of power 20-volts betweenNodes PWFM+and PWFM0 controller U400 generates an internal 5.0 referencevoltage to pin 7 node REF1. Output pin 6 node CLK is set highapproximately 20-volts (see oscillograph trace G6 segment 60 FIG. 34).C404 starts to charge through R401 until the voltage across C404 at pin3 reaches the comparator level (see oscillograph trace G1 segment 61FIG. 34) at resetting the pin 6 low (see oscillograph trace G6 segment62 FIG. 34). Capacitor C404 rapidly discharges though D401 (seeoscillograph trace G1 segment 63 FIG. 34). Pin 3 remains 0.6-volts abovePWFM0 node during the period pin 6 is low (see oscillograph trace G1segment 64 FIG. 34). On the rising edge of pin 6 capacitor C405 beginsto rapidly charge until the voltage in pin 4 reaches the internalcomparator level (see oscillograph trace G4 segment 65 FIG. 34). Thecomparator triggers internal transistor to rapidly discharge C404 (seeoscillograph trace G4 segment 66 FIG. 34). The cycle repeats with outputpin 6 being set high. External feedback current applied to U400 pin 1and node PW1 (see oscillograph trace G1 segment FIG. 34) follows theactual output voltage. Oscillograph trace G1 segment 67 (FIG. 34) is theperiod when the output switch conducting storing energy in the NSME.Oscillograph trace G1 segment 68 (FIG. 34) is the period when the outputswitch is off allowing storing energy in the NSME to be transferred tothe storage capacitor. Application of external current source or feedback network to pin 1 or node PW1 allows the pulse width to bemodulated. Removing current from PW1 lowers the comparator level causingthe comparator to trigger at lower voltages across C404 reducing thepulse width. Introducing current into node PW1 increases pulse widthfrom nominal to maximum of 95%. Resistor R404 and C404 determine thenominal pulse width. This design allows the CLK output to be pulse widthmodulated. Application of external feed back network to pin 4 or nodeFW1 allows the frequency to be modulated. Removing current from FW1slows the charging of C405. Longer charging time lowers the frequencyfrom the nominal setting. This arrangement allows the CLK output tofrequency modulated. When used with a resonant controller, R403 and C405determine the nominal frequency typically equal to the tank resonantfrequency. The external feedback is configured to lower the frequencyfrom nominal (maximum output) to zero frequency “off”. When used as apulse-width controller the nominal is set to maximum pulse width ofabout 90% feedback reduces the pulse-width. Sub-circuit PWFM may besimultaneously frequency and pulse width modulated. This configurationand mode of operation is unique to this instant invention. Feeding backof the output to the error amplifier is a unique mode of operation forcontrol element U400. Sub-circuit PWFM combines large dynamic range,precise control and fast response.

[0179]FIG. 34 Oscillograph traces of the PWFM (FIG. 33) controller inthe pulse-width modulation mode.

[0180]FIG. 35 Oscillograph trace of the TCTP (FIG. 8) resonant converterprimary voltage. FIG. 35 is an oscillograph trace of the voltagedeveloped across capacitor C10 (FIG. 8). In this embodiment the supplyVBAT was only 18-volts. The primary 100 (FIG. 18) inductance 203 uH wasachieved by 55 turns on a 26 u 2.28 oz. KoolMu magnetic element 101. Thesecondary winding 103 (FIG. 18) is 15 turns on core 101. A 5.5-watt loadis connected to winding 103. The NSME primary 100 (FIG. 18) developed anexcitation voltage of 229 volts peak more than 10 times VBAT. Tankconverters TCTP and TCSSC (FIG. 7) take advantage of the desirableproperties of the non-saturating magnetic to develop large flux biases.The useful large flux may harvested into useful power by addition of“flux nets” windings to the magnetic element.

[0181]FIG. 36 Regulated 18_Volt DC control power sub-circuit REGSub-circuit REG consists of resistor R517, regulator Q514 and capacitorsC514, C515, C516, C518, and C517. TABLE FIG. 36 Element Value/partnumber Q514 LM7818 C515 0.01 uF C517 0.01 uF C514 10 uF C518 10 uF

[0182] Pin REG0 connects to the external power source return. Node REG0is also the return line it connects to Q514 pin 2, and capacitors C518,C514, C515, and C517. Resistor R517 is connected to the pin 1 (input)node of voltage regulator Q514 and to input pin RIN+. Voltage regulatorQ514 Pin 3 is the 18 vdc regulated DC output is connected to thecapacitors C515, C514 and output pin 18V. Capacitors C515, C517 aresolid dielectric type is used to filter high frequency ripple and toprevent Q514 from oscillating. Sub-circuit REG provides regulated powerfor control circuits and output switch buffer AMP (FIG. 29).

[0183]FIG. 37 is a schematic for a non-isolated high side switch buckconverter sub-circuit HSBK. FIG. 37 is a non-isolated high side switchbuck converter sub-circuit HSBK. This converter topology consists of anon-isolated high efficiency buck stage, which provides regulated powerto an efficient push-pull isolation stage. Sub-circuit HSBK consists ofdiode D8, capacitor C8, FET transistor Q31, sub-circuit TCTP (FIG. 8),sub-circuit BL1, (FIG. 18B), sub-circuit IFB (FIG. 40B), sub-circuit AMP(FIG. 29) and sub-circuit PWFM (FIG. 33). TABLE FIG. 37 ElementValue/part number C68 250 uf D68 MUR820 Q31 IRF540N

[0184] External power source VBAT connects to pins DCIN+ and DCIN−. PinDCIN+ connects to transistor Q31 source, sub-circuit PWFM pin PWFM0,sub-circuit AMP pin GA0, and sub-circuit IFB pin FBE, sub-circuit TCTPpins DCIN+ and B−. Regulated 18-volt output from sub-circuit TCTP pin B+connects to sub-circuit AMP pin GA+and to sub-circuit PWFM pin PWFM+.This provides the positive gate drive relative to the source of Q31.Power source VBAT return is connected to pin DCIN−, sub-circuit TCTP pinDCIN−, diode D68 anode, capacitor C68, RLOAD, sub-circuit IFB pin OUT−,output pin B− and ground/return node GND. Sub-circuit PWFM is designedfor adjustable pulse-width operation from 0 to 90%, maximum pulse widthoccurs with no feedback current to pin PW1. Increasing the feedbackcurrent reduces the pulse-width and output voltage from converter HSBK.Sub-circuit PWFM clock/PWM output pin CLK is connected to the input pinGA1 of buffer sub-circuit AMP. The output of sub-circuit AMP pin GA2 isconnected to the gate of Q31. The drain of Q31 is connected tosub-circuit BL1 pin P1B and the cathode of D68. Pin P1A of sub-circuitBL1, is connected to capacitor C8, sub-circuit IFB pin OUT− and RLOAD.With sub-circuit PWFM pin CLK high buffer AMP output pin GA2 charges thegate of transistor switch Q31. Switch Q31 conducts charging capacitorC68 through NSME BL1 from source VBAT and storing energy in BL1.Feedback output pin FBC from sub-circuit IFB is connected to sub-circuitPWFM pulse-width adjustment pin PW1. As the output voltage reaches thedesigned level sub-circuit IFB removes current from PWl commanding PWFMto reduce the pulse-width or on time of signal CLK. After sub-circuitPWFM reaches the commanded pulse-width PWFM switches output pin CLK lowturning off Q31 stopping the current into BL1. The stored energy isreleased from NSME BL1, into the now forward biased diode D68 chargingcapacitor C68. By modulating the on time of switch Q31 the converter“bucks” applied voltage and efficiently regulates to a lower voltage.Regulated voltage is developed across Nodes B− and B+. Sub-circuit IFBprovides the isolated feedback voltage to the sub-circuit PWFM. Whensub-circuit IFB senses the converter output (nodes B+ and B−) is at thedesigned voltage more current is conducted by the phototransistor.Sinking current from PM1 commands the PWFM to a shorter pulse-width thusreducing the converter output voltage. In the event the feedback signalfrom IFB commands the PWFM to minimum output. Gate drive to switch Q31is removed stopping all buck activity capacitor C68 discharges throughRLOAD. Input current from VBAT is sinusoidal making the converter veryquiet. As such the switch Q31 is not exposed to large current spikescommon to saturating magnetic prior art. Thus placing less stress on theswitches thereby increasing the MTBF. Sub-circuit HSBK takes advantageof the desirable properties of the NSME in this converter topology.

[0185]FIG. 38 is a schematic for an isolated two-stage low side switchbuck converter sub-circuit LSBKPP. This converter topology consists of ahigh efficiency low-side switch buck stage, which provides regulatedpower to an efficient push-pull isolation stage. An efficient center-tapfullwave rectifier provides rectification. Sub-circuit LSBKPP consistsof diode D46, capacitor C46, FET transistor Q141, sub-circuit REG (FIG.36), sub-circuit OUTB (FIG. 25A), sub-circuit BL1, (FIG. 18B),sub-circuit TCTP (FIG. 8), sub-circuit IFB (FIG. 40B), sub-circuit AMP(FIG. 29), sub-circuit DCAC1, and sub-circuit PWFM (FIG. 33). TABLE FIG.38 Element Value/part number C46 250 uf D46 MUR820 Q141 IRFS40N

[0186] External power source VBAT connects to pins DCIN+ and DCIN−. Frompin DCIN+ connects to sub-circuit REG pin RIN+, D46 cathode, capacitorC46, sub-circuit TCTP (FIG. 8) pin DCIN+, and sub-circuit DCAC1 pin DC+.Voltage regulator sub-circuit REG output pin +18V connects tosub-circuit AMP pin GA+ and to sub-circuit PWFM pin PWFM+. Sub-circuitREG provides regulated low voltage power to the controller and to themain switch buffer. VBAT negative is connected to pin DCIN− and groundreturn node GND. Node GND connects to sub-circuit PWFM pin PWFM0,sub-circuit AMP pin GA0, Q141 source, sub-circuit IFB pin FBE,sub-circuit REG pin REG0 and sub-circuit TCTP pin DCIN−. Sub-circuitPWFM (FIG. 33) is designed for variable pulse width operation. Thenominal frequency is between 20-600 Khz PWFM is configured for maximumpulse width 90% (maximum buck voltage) with no feedback current fromsub-circuit IFB. Increasing the feedback current reduces the Q111 ontime reducing the voltage to the push-pull stage and the output fromconverter LSBKPP. Sub-circuit PWFM clock output pin CLK is connected tothe input pin GA1 of buffer sub-circuit AMP (FIG. 29). The output ofswitch speed up buffer sub-circuit AMP pin GA2 is connected to the gateof Q141. Floating isolated 18-volt power from sub-circuit TCTP pin B+connects to sub-circuit DCAC1 pin P18V. The drain of Q141 is connectedto sub-circuit BL1, pin P1A and the anode of D46. The return line ofsub-circuit DCAC1 pin DC− connects to sub-circuit BL1, pin P1B,sub-circuit TCTP pin B− and C46. With sub-circuit PWFM pin CLK highbuffer AMP output pin GA2 charges the gate of transistor switch Q141.Switch Q141 conducts reverse biasing diode D46; capacitor C46 startscharging through NSME BL1, from source VBAT. During the time Q141 isconducting, energy is stored in NSME sub-circuit BL1. Charging C46provides power to final push-pull converter stage DCAC1. The output ofthe output rectifier sub-circuit OUTB is connected to feedbacksub-circuit IFB output pin FBC from sub-circuit IFB is connected tosub-circuit PWFM pulse-width adjustment pin PW1. Sub-circuit IFB removescurrent from PW1 commanding PWFM to reduce the pulse-width or on time ofsignal CLK. After sub-circuit PWFM reaches the commanded pulse-widthPFFM switches CLK low turning off Q141 stopping the current into BL1.The energy is released from NSME BL1, into the now forward biasedflyback diode D46 charging capacitor C46. By modulating the on time ofswitch Q141 the converter voltage is regulated. Regulated voltage isdeveloped across C46 Nodes DC+ and GND. Providing energy to the isolatedconstant frequency push-pull DC to AC converter sub-circuit DCAC1 (FIG.2). Sub-circuit DCAC1 provides efficient conversion of the regulatedbuck voltage to a higher or lower voltage set by the magnetic elementwinding sub-circuit PPT1 (FIG. 19) ratio. The center tap of thepush-pull output magnetic is connected to, sub-circuit OUTB pin OUT−,RLOAD, sub-circuit IFB pin OUT− and the pin OUT− forming the return linefor the load and feedback network. Output of sub-circuit DCAC1 pin ACHis connected to sub-circuit OUTB pin C7B. Output of sub-circuit DCAC1pin ACL is connected to sub-circuit OUTB pin C8B. Sub-circuit OUTBprovides rectification of the AC power generated by sub-circuit DCAC1.As the non-saturation magnetic converter is very quite minimal filteringis required by OUTB. This further reduces cost and improves efficiencyas losses to filter components are minimized. Sub-circuit IFB providesthe isolated feedback current to the sub-circuit PWFM. When sub-circuitIFB senses the converter output (nodes OUT+ and OUT−) is greater thanthe designed/desired voltage, current is removed from node PM1. Sinkingcurrent from PM1 commands the PWFM to a shorter pulse-width thusincreasing the buck action and reducing the first stage converter outputvoltage. In the event the feedback signal from IFB commands the PWFM tominimum output. Gate drive to switch Q141 is removed stopping all buckactivity capacitor discharging C46. Input current from VBAT to chargeC46 is sinusoidal making the converter very quiet. In addition theswitch Q141 is not exposed a potentially destructive current spike.Placing less stress on the switches thereby increasing the MTBF.Sub-circuit LSBKPP takes advantage of the desirable properties of theNSME in this converter topology. Adjusting the NSME BL1, (FIG. 18B) setsthe amount of buck voltage available to the final push-pull isolationstage. Greater efficiencies are achieved at higher voltages. The finaloutput voltage is set by the turns ratio of the push-pull element PPT1(FIG. 19). Converter LSBKPP provides efficient conversion from highvoltage sources into high current isolated output.

[0187]FIG. 39 is a schematic for an isolated two-stage low side switchbuck converter sub-circuit LSBKPPBR. This converter topology consists ofa non-isolated high efficiency low-side switch buck stage, whichprovides regulated power to an efficient push-pull isolation stage. Afullwave bridge rectifier provides rectification. Sub-circuit LSBKPPBRconsists of diode D6, capacitor C6, FET transistor Q111, sub-circuit REG(FIG. 36), sub-circuit OUTBB (FIG. 25B), sub-circuit BL1, (FIG. 18B),sub-circuit TCTP (FIG. 8), sub-circuit IFB (FIG. 40B), sub-circuit AMP(FIG. 29), sub-circuit DCAC1 (FIG. 2), and sub-circuit PWFM (FIG. 33).TABLE FIG. 39 Element Value/part number C6 250 uf D6 MUR820 Q111 IRFP

[0188] External power source VBAT connects to pins DCIN+ and DCIN−. Frompin DCIN+ connects to sub-circuit REG pin RIN+, D6 cathode, capacitorC6, sub-circuit TCTP (FIG. 8) pin DCIN+, and sub-circuit DCAC1 pin DC+.Voltage regulator sub-circuit REG output pin +18V connects tosub-circuit AMP pin GA+ and to sub-circuit PWFM pin PWFM+. Sub-circuitREG provides regulated low voltage power to the controller and to themain switch buffer. VBAT negative is connected to pin DCIN− connects tosub-circuit PWFM pin PWFM0, sub-circuit AMP pin GA0, Q111 source,sub-circuit IFB pin FBE, sub-circuit REG pin REG0, sub-circuit TCTP pinDCIN−. Sub-circuit PWFM (FIG. 33) is designed for variable pulse widthoperation. The nominal frequency is between 20-600 Khz PWFM isconfigured for maximum pulse width 90% (maximum buck voltage) with nofeedback current from sub-circuit IFB. Increasing the feedback currentreduces the Q111 on time reducing the voltage to the push-pull stage andthe output from converter LSBKPPBR. Sub-circuit PWFM clock output pinCLK is connected to the input pin GA1 of buffer sub-circuit AMP (FIG.29). The output of switch speed up buffer sub-circuit AMP pin GA2 isconnected to the gate of Q111. Floating isolated 18-volt power fromsub-circuit TCTP pin B+ connects to sub-circuit DCAC1 pin P18V. Thedrain of Q111 is connected to sub-circuit BL1 pin PA1 and the anode ofD6. The return line of sub-circuit DCAC1 pin DC-connects to sub-circuitBL1, pin P1B, sub-circuit TCTP pin B− and C6. With sub-circuit PWFM pinCLK high buffer AMP output pin GA2 charges the gate of transistor switchQ111. Switch Q111 conducts reverse biasing diode D6; capacitor C6 startscharging through NSME BL1 from source VBAT. During the time Q111 isconducting, energy is stored in NSME sub-circuit BL1. Charging C6provides power to final push-pull converter stage DCAC1. The output ofthe output rectifier sub-circuit OUTBB is connected to feedbacksub-circuit IPB output pin FBC from sub-circuit IFB is connected tosub-circuit PWFM pulse-width adjustment pin PW1. Sub-circuit IFB removescurrent from PWl commanding PWFM to reduce the pulse-width or on time ofsignal CLK. After sub-circuit PWFM reaches the commanded pulse-widthPFFM switches CLK low turning off Q111 stopping the current into BL1.The energy is released from NSME BL1 into the now forward biased flybackdiode D6 charging capacitor C6. By modulating the on time of switch Q111the converter voltage is regulated. Regulated voltage is developedacross C6 nodes DC+ and DC−. Providing energy to the isolated constantfrequency push-pull DC to AC converter sub-circuit DCAC1 (FIG. 2).Sub-circuit DCAC1 provides efficient conversion of the regulated buckvoltage to a higher or lower voltage set by the magnetic element windingsub-circuit PPT1 (FIG. 19) ratio. The return node of the sub-circuitOUTBB pin OUT− is connected to RLOAD, sub-circuit DCAC1 pin AC0,sub-circuit IFB pin OUT− and the pin OUT−. Node OUT− is the return linefor the load and feedback network. Output of sub-circuit DCAC1 pin ACHis connected to sub-circuit OUTBB pin C7B. Output of sub-circuit DCAC1pin ACL is connected to sub-circuit OUTBB pin C8B. Sub-circuit OUTBBprovides rectification of the AC power generated by sub-circuit DCAC1.As the disclosed non-saturation magnetic converter has minimal outputripple, less filtering is required by OUTBB. This further reduces costand improves efficiency as losses in filter components are minimized.Sub-circuit IFB provides the isolated feedback current to thesub-circuit PWFM. Open collector output of IFB pin FBC connects to PWFMpin PW1. When sub-circuit IFB senses the converter output (nodes OUT+and OUT−) is greater than the designed/desired voltage, current isremoved from node PM1. Sinking current from PM1 commands the PWFM to ashorter pulse-width thus increasing the buck action and reducing thefirst stage converter output voltage. In the event the feedback signalfrom IFB commands the PWFM to minimum output. Gate drive to switch Q111is removed stopping all buck activity capacitor discharging C6. As theNSME does not saturate the destructive noisy current spikes common toprior art are absent. Input current from VBAT to charge C6 is sinusoidalmaking the converter very quiet. In addition the switch Q111 is notexposed a potentially destructive current spike. Placing less stress onthe switches thereby increasing the MTBF. Sub-circuit LSBKPPBR takesadvantage of the desirable properties of the NSME in this convertertopology. Adjusting the NSME BL1, (FIG. 18B) sets the amount of buckvoltage available to the final push-pull isolation stage. Greaterefficiencies are achieved at higher voltages. The final output voltageis set by the turns ratio of the push-pull element PPT1 (FIG. 19).Converter LSBKPPBR provides efficient conversion from high voltagesources such as high power factor AC to DC converters such assub-circuit ACDCPF (FIG. 4).

[0189]FIG. 40 PFC over voltage feed back sub-circuit IPFFB

[0190]FIG. 40 is the schematic of the inventions isolated over voltagefeed back network sub-circuit IPFFB. Sub-circuit IPFFB consists ofResistors R926, R927, R928, R929 and R930, capacitor C927, zener diodesD928 and D903, transistor Q915 and opto-isolator U903. TABLE FIG. 40Element Value/part number U903 NEC2501 Q915 FZT705CT D903 ML5248B (18 v)D928 1SMB5956BT3 (200 v) R926 20k ohms R927 10k ohms R928 10k ohms R92910k ohms R930 20k ohms

[0191] Node PF+ connects through resistor R927 to cathode of D903 andanode of opto-isolator U903. Cathode of diode D903 is connected to pinPF+. Resistor R928 is connected from anode of D928 to base of Q915.Capacitor C927 is connected in parallel with zener diode D903. ResistorR928 limits maximum base current. Resistor R929 is connected betweenbase and emitter of Q915. Resistor R929 is used to shunt excess zenerleakage current from the base common in high voltage diodes. Twohundred-volt zener diode cathodes D928 are connected to pin PF+. Anodeof D928 is connected to R930 and R928. Resistor R930 provides a path forleakage current from 200-volt zener D928. Resistor R926 limits themaximum current to U903 internal light emitting diode to about 10 ma.Resistor R927 sets the maximum zener current at maximum boost voltage ofapproximately 200-volts to 20 ma. Transistor Q915 is biased off when thevoltage from node PF+ and PF− is less than the zener voltage of200-volts. Transistor is in a cutoff or non-conducting state no currentis injected to U903 LED. The internal phototransistor is also in anon-conducting state. The attached external control sub-circuit is notcommanded to change its output. With 200 volts or more applied to nodesPF+ and PF− reverse biased zener diode D928 injects current into thebase of Q915. Resistor R927, capacitor C927 and diode D903 provide18-volts to the collector of Q915. Transistor Q915 conducts current intoU903 LED injecting base current into the U903 phototransistor.Modulating the LED current is reflected as variable impedance betweenFBC and FBE. This phototransistor may be connected as a variable currentsource or impedance. This sub-circuit senses excessive boost voltage andquickly feeds back to the control sub-circuit (See PFA (FIG. 23), PFB(FIG. 24) or (PWFM FIG. 33)) automatically reducing the boost voltage.

[0192]FIG. 40A is a schematic diagram of the non-isolated boost outputvoltage feed back sub-circuit FBA. Sub-circuit FBA consists of ResistorsR1120, R1121, R1122, R1123 and R1124. TABLE FIG. 40A Element Value/partnumber R1123 499k ohms R1124 499k ohms R1122 6.65k ohms R1121 499k ohmsR1120 1 MEG ohms

[0193] Input node PF+ connected to series resistor [R1123+R1124] then toparallel resistors [R20∥R21∥R22] to the return node BR−. ResistorsR1120, R1121, R1122, R1123 and R1124 values are selected for a nominalinput voltage of 385-volts and output feed back voltage of 3.85. (Seeoscillograph G1 FIG. 34) Resistors R1120, R1121, R1122, R1123 and R1124are shown in surface mount configuration but can be combined into twothru hole-resistors. Feedback output node PF1 is connected to node PF1of sub-circuit PFA (FIG. 23) or PFB (FIG. 24). Return pin BR− isconnected to BR− of PFA (FIG. 23) or PFB (FIG. 24). Nodes FBE and FBC itmay also be connected between nodes FM1 pin PWFM0 or PW1 pin PWFM0 ofcontrol sub-circuit PWFM (FIG. 33).

[0194]FIG. 40B output voltage feed back sub-circuit IFB

[0195]FIG. 40B is the schematic of the inventions isolated low voltagefeed back network sub-circuit FBA. Sub-circuit IFB consists of ResistorsR900, R901 and R902, zener diode D900, Darlington transistor Q900 andopto-isolator U900. TABLE FIG. 40B Element Value/part number U900NEC2501 Q900 FZT705CT D900 1N5261BDICT R900  1k ohms R902  4k ohms R90140k ohms

[0196] Node OUT+ connects cathode of D900 to R901. Anode of diode D900is connected to series resistor R900 to base of Darlington transistorQ900. Resistor R902 is connected from base to emitter to Q900. ResistorR901 connects to anode of opto-isolator U900 LED (light emitting diode)the cathode is connected to Q900 collector. Emitter of Q900 is thereturn current path and connects to pin/node OUT−. Resistor R901 limitsthe maximum current to U900 internal light emitting diode to 20 ma.Resistor R902 shunts some of the zener leakage current from the base.Zener diode voltage selection sets the converter output voltage atypical value maybe 48-volts. The zener voltage is the final desiredoutput minus two base emitter junction drops (1.4V). Once the OUT+ nodereaches the zener voltage a small base current biases Q900 into aconducting state turning“on” opto-isolator U900 internal LED. ResistorR900 limits the maximum base current to Q900. Resistors R900 and R901are selected to bias Darlington transistor Q900 collector current withnominal voltage across nodes OUT+ and OUT−. Change in voltage betweenOUT+ and OUT− modulates the opto-isolator U900 LED current in turnchanging the base current of U900 internal photo transistor.Phototransistor emitter is node FBE collector is node FBC. Modulatingthe LED current is reflected as variable impedance between FBC and FBE.This phototransistor may be connected as a variable current source orimpedance. When used with control sub-circuit PFA (FIG. 23), PFB (FIG.24) or (PWFM FIG. 33) the phototransistor is connected as a currentshunt. Higher voltage applied to OUT+ and OUT− nodes increases thefeedback shunt current commanding the control sub-circuit (See PFA (FIG.23) or PFB (FIG. 24) or PWFM (FIG. 33)) to reduce the pulse-width orfrequency. IFB accomplishes high speed feed back due to the very highgain of the Darlington transistor and the rapid response of the internalconverter stage(s) active ripple reduction and excellent load regulationare achieved.

[0197]FIG. 40C is the schematic of the alternate PFC isolated overvoltage feed back network sub-circuit IOVFB. Sub-circuit IOVFB consistsof resistors of R917, R938, R939 and R940, diode D911, Darlingtontransistor Q914 and opto-isolator U905. TABLE FIG. 40C ElementValue/part number U905 NEC2501 Q914 FZT705CT R938 160k ohms R939  70kohms D911 1N5261BOTCT R940  50k ohms R917  40k ohms

[0198] The output of the PFC at pin PF+ is connected to R917 then tocollector of Q914. Resistor R917 sets the maximum current to U905 lightemitting diode. Resistor R938 is connected from return node PF+ to zenerdiode D911 cathode and R938. Resistor R939 is connected from return nodePF− to zener diode D911 cathode and R938. Anode of D911 is connected towiper arm of adjustable resistor R940. One leg of R940 is connected tothe base of transistor Q914 the other to R939 and U905 LED anode andR939. Emitter of Q914 is connected to anode of U904. Adjustable resistorR940 sets the maximum or trip voltage before transistor Q914 is biasedon. Providing current to U905 LED. Phototransistor emitter is node FBEcollector is node FBC. Modulating the LED current is reflected asvariable impedance between FBC and FBE. This phototransistor is normallyconnected as a shunt to force the control element to a minimum output.This sub-circuit senses the boost voltage and feeds back to the PFC.Where excessive boost voltage forces the PFC to automatically reduce theboost voltage.

[0199]FIG. 41 output voltage feed back sub-circuit FBI

[0200]FIG. 41 is the schematic of the alternate low voltage feed backnetwork sub-circuit FBI. Sub-circuit FBI consists of Resistors R81, R82and R83, zener diode D80, NPN transistor Q80 and capacitor C80. TableElement Value/part number R81 1 k ohms D80 Zener Voltage = (DesiredOutput-0.65 V) Q80 BCX70KCT C80 1000 pf R82 1 k ohms R83 715 k ohms

[0201] Node OUT+ connects cathode of D80. Anode of diode D80 isconnected to through resistor R83 to OUT− and resistor R82 to base oftransistor Q80. Capacitor C80 is connected from base to pin OUT−.Capacitor C80 bypasses high frequency to noise to OUT−. Resistor R81 isconnected from emitter of Q80 to node OUT−. Resistor R81 adds localnegative feedback to reduces the effects of variation in transistorgain. Collector of Q80 is connected to pin FBC. The return current nodeconnects to pins FBE and OUT−. Resistor R82 limits the maximum basecurrent protecting Q80. Resistor R83 shunts some of the zener leakagecurrent from the base. Zener diode voltage selection sets the converteroutput voltage a typical value maybe 48-volts. The zener voltage is thefinal desired output minus one base emitter junction drop (0.65-Volts).When the OUT+ node reaches the nominal level reverse biased zener startsto conduct injecting a small base current into Q80. Biasing transistorinto a conducting state. Change in voltage between OUT+ and OUT−modulates Q80 collector current. During normal operation the zener diodeis biased at it's knee thus small changes in voltage result inrelatively large collector current changes. When sub-circuit FBI usedwith control sub-circuit PFA (FIG. 23), PFB (FIG. 24) or (FIG. 33) thetransistor is connected as a current shunt. Higher voltage applied toOUT+ and OUT− nodes increases the feedback shunt current commanding thecontrol sub-circuit (See PFA (FIG. 23) or PFB (FIG. 24) or PWFM (FIG.33) to reduce the pulse-width or frequency. Sub-circuit FBI provideshigh-speed feedback and gain to ripple components. With the rapidresponse of the internal converter stage(s) active ripple reduction andexcellent load regulation are achieved.

[0202]FIG. 42 over voltage protection sub-circuit OVP1

[0203]FIG. 42 is the schematic of the inventions over voltage protectionembodiment sub-circuit OVP1. Sub-circuit OVP1 consists of SCR (siliconcontrolled rectifier) SCR1200, resistor R1200, capacitor C1200 and zenerdiodes D1200, D1202 and D1203. Table Element Value/part number SCR1200MCR265-10 D1203 BZT03-C200 (200 V) D1202 BZT03-C200 (200 V) D1200 IN4753(5.1 v) C1200 220 pf R1200 10, 0k ohms

[0204] Input pin PF+ is connected to cathode of zener diode D1203, anodeof D1203 is connected to series zener diodes [D1202+D1200] then to gateof SCR1200. Noise attenuation network of [R1200∥C1200] is connected fromSCR SCR1200 gate to the return node BR−. Diodes D1102 and D1103 are both200-volt; D1101 is a 5.1-volt type the sum of the zener voltages set thetrip point of the OVP at 405-volts. Other trip voltages may beimplemented by selecting other zener diode combinations. Capacitor C1200and R1200 prevents leakage current and transients from accidentallytripping the OVP. In the event of very high AC line voltages or acomponent failure in a feed back loop (FIG. 40A, 40B, 40C. or 40) Theboost voltage may quickly rise increase to levels dangerous to theoutput switch or output storage capacitors. When the output boostvoltage of the at node PF+ rises above 405V, zener diodes D1203, D1202and D1200 conduct a small current into the gate of SCR1200 turningSCR1200 on. Turning SCR1200 on places a low impedance path across the ACline through the rectifier sub-circuit BR (FIG. 22). SCR1200 and bridgerectifier diodes must be selected to withstand the short circuitcurrents that may exceed 100 amperes until the input fuse opens. Thusquickly limiting the boost output voltage to a safe level. This circuitshould never operate under normal AC line voltages. By changing zenervoltages this sub-circuit would also be suitable for use in the acrossthe rectifier output to protect the load from an over voltage condition.Sub-circuits OVP1 shuts down the converter with out opening the linefuse. Sub-circuit OVP may be used in combination with OVP1 (FIG. 42A) asa fail-safe back up for critical loads.

[0205]FIG. 42A over voltage protection sub-circuit OVP1

[0206]FIG. 42A is the schematic of the inventions over voltageprotection embodiment sub-circuit OVP2. Sub-circuit OVP2 consists ofSCRs (silicon controlled rectifier) SCR1101 and SCR1100, resistors R1101and R1102, capacitors C1100 and C1101 and zener diodes D1100, D1102 andD1103. Table Element Value/part number SCR1101 S101E (Teccor) SCR1100S601E (Teccor) D1103 BZT03-C200 (200 V) D1102 BZT03-C200 (200 V) D1100IN4753 (5.1 v) R1100 16000 R1101 5.1 K ohms R1102 5.1 K ohms C1100 1200pf C1101 1200 pf

[0207] Anode of SCR1101 is node/pin CP18V+ that is connected to externalcontrol DC source. Return node BR− is connected to SCR1101 cathode andcapacitor C1100. Input node PF+ is connected to cathode of zener diodeD1103 and to series resistor R1100 then to anode of SCR SCR1102. Theanode of D1103 is connected to the cathode of D1102. The anode of D1102is connected to the cathode of D1100. The cathode of SCR1100 isconnected to the gate of SCR1101. The anode of D1103 is connected toseries zener diodes [D1102 +D1100] then to capacitor C1100 then to thereturn node BR−. Capacitor [C1200∥R1200] prevents leakage current andtransients from accidentally tripping OVPB. In the event of very high ACline voltages or a component failure in a feed back loop (IPFFB FIG.40A, FBA 40B, IFB 40C. or FBI FIG. 41) The boost voltage may quicklyrise increase to levels dangerous to the output switch or output storagecapacitors. When the output boost voltage of the at node PF+ rises above405V, zener diodes D1103, D1102 and D1100 conduct a small current intothe gate of SCR1101 latching SCR1101 on. Resistor R1100 provides holdingcurrent for SCR1101. Turning SCR1101 provides gate current to SCR1100,resistors R1100 and R1101 limits the gate current and provides the holdcurrent to SCR1100. With gate current to SCR1100 the SCR is turned onproviding a low impedance path from nodes CP18V+ to BR−. This actionremoves the regulated power to the main switch buffer and or PWMcontrollers PFA (FIG. 23) or PWFM (FIG. 33) and or buffer AMP (FIG. 29)thus turning off the main switch. The converter is held in an off stateuntil boost voltage PF+ through R1100 can not maintain the holdingcurrent of SCR1101. Typically power must be removed from the system toreset SCR1101. The minimum holding current of SCR1101 is typically 5-10ma. The action of OVP1 quickly limits the boost output voltage to a safelevel. This circuit should never operate under normal AC line voltages.By changing zener voltages this sub-circuit would also be suitable foruse across the output rectifier to protect the load from an over voltagecondition. Sub-circuit OVP1 gracefully shuts down the converterrequiring manual intervention to reset the fault.

[0208]FIG. 42B is the schematic of the isolated output over voltage feedback network sub-circuit OVP2. Sub-circuit OVP2 consists of resistors ofR970, R971, and R972, capacitor C970, zener diode D970, SCR SCR970,Darlington transistor Q970 and opto-isolator U970. Table ElementValue/part number D970 1N5261BOTCT U970 NEC2501 Q970 FZT705CT R970 160 kohms R971 10 k ohms R972 22 k ohms C970 200 pf

[0209] The output of the converter at pin OUT+ is connected to R972 andto the cathode of zener diode D970. The anode of D970 is connected toseries resistor R970 then to base of Q970. Resistor R970 sets themaximum base current to Q970. Resistor R971 is connected between theanode of D970 and return node OUT− Anode of light emitting diode U970 isconnected to resistor R972 then to OUT+. The cathode of U970 LED isconnected to Q980 collector. Emitter of Q980 is connected to return nodeOUT−. Zener diode D960 sets the maximum or trip voltage beforetransistor Q970 is biased on providing current to U970 LED. Applicationof voltage greater than the zener voltage of D970 injects a small basecurrent into Q970. Transistor Q970 turns on the internal LED of U970placing phototransistor in a conducting state and low impedance to pinsOVC and OVC. External push-pull driver sub-circuit PPG (FIG. 43) is shutdown immediately by bringing pin PPEN high stopping the output stage.Sub-circuit OVP2 senses the output voltage and quickly feeds back to thepush-pull PFC. Where excessive boost voltage forces the PFC toautomatically reduce the boost voltage.

[0210]FIG. 42C is the schematic of the isolated output over voltagecrowbar network sub-circuit OVP3. Sub-circuit OVP3 consists of resistorsof R980, R981, R982, R983, R984 and R985, capacitors C980, C981 and C982zener diode D980, SCRs SCR980 and SCR981, Darlington transistor Q980 andopto-isolator U980. Table Element Value/part number D980 1N5261BOTCTSR980 S601E (Teccor) U980 NEC2501 Q980 FZT705CT R980 160 k ohms R981 10k ohms R982 22 k ohms R983 51 k ohms R984 1200 ohms R985 510 ohms C980200 pf C981 1200 pf C982 1200 pff

[0211] The converter output is sensed at pin OUT+ reference to pin OUT−.Pin OUT+ is connected to resistor R982 and to the cathode of zener diodeD980. The anode of D980 is connected to series resistor R980 then tobase of Q980. Resistor R980 limits the base current to Q980. ResistorR981 is connected between the anode of D980 and return node OUT− toprovide a diode leakage current path. Anode of light emitting diode U980is connected through resistor R982 then to OUT+. The cathode of U980 LEDis connected to Q980 collector. Emitter of Q980 is connected to returnnode OUT−. Zener diode D960 sets the maximum or trip voltage beforetransistor Q980 is biased on providing current to U980 LED. Applicationof voltage greater than the zener voltage of D980 injects a small basecurrent into Q980.Emitter of opto-isolator U980 is connected to the gateof SCR981 and through [R984∥C982] to return node BR−. Transistor Q980turns on the internal LED of U980 placing phototransistor in aconducting state and supplying gate current to SRC SCR981 from theexternal 18-volt source connected to pin CP18V+. Network [R984∥C982]prevents false triggering of SCR SCR981. The cathode of SCR SCR981 isconnected to the gate of SCR SCR980 and through [R985∥C981] to returnBR−. With SCR SCR981 turned on gate current is provided to low voltageSCR SCR980. High voltage boost output is connected to pin PF+ resistorR983 supplies hold current to SCR SCR981 holding SCR SCR980 on. SCRSCR980 is selected for low hold current and ability to block the maximumboost voltage on PF+. SCR SRC980 anode is connected to pin CP18V+. SCRSRC980 cathode is connected to return pin BR−. SCR980 clamps the lowvoltage supply CP (FIG. 26) or CPA (FIG. 27). With the low supply helddown the gate drive to the main switch is disabled turning off theconverter. With the main switch Q1 (FIG. 1,3,4) turned off holdupcapacitor C17 charges to applied AC line peak. With pin PF+ held nearline peak SCRs SCR981 will hold SCR SCR981 on until AC line power isremoved to the converter. Sub-circuit OVP3 senses the out ofspecification output voltage and quickly stop the converter therebyprotecting the load and converter with out generating destructivecurrents like OVP (FIG. 42).

[0212]FIG. 43 Push-pull oscillator sub-circuit PPG FIG. 43 is thepush-pull oscillator sub-circuit of the invention. The currentimplementation uses a Motorola MC33025 pulse width modulator IC togenerate the clock signals to drive the push-pull output stage.Sub-circuit PPG consists of U14 a two-phase oscillator, resistors R126,R130, R131, R132, R133, R134, R135, R136 and R137, capacitors C143,C136, C139, C140, C141 and C142. Table Element Value/part number U14MC33025 R126 12 k ohms R130 10 ohms R131 10 ohms R132 47 k ohms R133 10k ohms R134 100 k ohms R135 15 k ohms R136 1.5 MEG ohms R137 15 k ohmsC136 0.22 uf C139 0.22 uf C140 0.22 uf C141 0.01 uf C142 0.001 uf C143.33 uf

[0213] The current implementation uses a Motorola MC33025 pulse widthmodulator IC to generate the clock signals to drive the push-pull stage.But, any non-overlapping two phase fixed frequency generator could beused. Pin 1 of U14 is connected to [capacitor C143∥Resistor R132] thento pin 3. Resistor R134 connects the internal 5.1-volt reference outputof U14 pin 16 to pin 1. Resistors R135 in series with R137 from 5.1-voltreference to return node PPG0 form a voltage divider; the center isconnected to U14 pin 2 placing pin 2 at 2.55-volts. Resistor R126 isconnected from U14 pin 5 to return node PPG0. Resistor R133 is connectedfrom U14 pin 1 to return node PPG0. Timing capacitor C142 is connectedfrom U14 pins 6 and 7 to return node PPG0. Resistor R126 and capacitorC142 set the operating frequency of the internal oscillator. Timingresistor could be replaced with a JFET, MOSFET, transistor, or similarswitching device to provide variable frequency operation. The drain ofthe transistor would be connected to pin 5. The source would beconnected to return node PPG0. The variable frequency commandvoltage/current is applied between gate and source. Capacitor C141 isconnected from U14 pin 8 to return node PPG0. Capacitor C136 isconnected from U14 pin 16 to return node PPG0. Capacitor C140 isconnected from U14 pin 15 to return node PPG0. Capacitor C139 isconnected from U14 pin 13 to return node PPG0. Resistor R136 isconnected from U14 pin 9 to return node PPG0. U14 pins 10 and 12 isconnected to return node PPG0. External power is connected to node/pinPPG+ to PWM (pulse width modulator) IC U14 on pin 15 through resistorR130 connected to the 18-volt control supply. Resistor R131 connected topin 13 of U14 and PPG+ provides power to the totem-poll output stage.The power return line is connected to node PPG0. IC U14 is designed tooperate at a constant frequency of approximately 20.-600 Khz with afixed duty cycle of 35-49.9%. Resistors R135, R137, R133 configure U14to operate at maximum pulse width. A two-phase non-over lapping squarewave is generated on pins 11 node PH2 and pin 14 node PH1 and deliveredto speed-up buffers AMP described in FIG. 29. The two-phase generator isconfigured to prevent the issue of overlapping drive signals that wouldnull the core bias and present excessive current to the switches.Sub-circuit PPG provides the drive to the push-pull switches makingefficient use of the NSME. Although the present invention has beendescribed with reference to a preferred embodiment, numerousmodifications and variations can be made and still the result will comewithin the scope of the invention. No limitation with respect to thespecific embodiments disclosed herein is intended or should be inferred.

We claim:
 1. A transformer for use in a converter of a power supply,said transformer comprising: a core; a primary winding wrapped aroundthe core; at least one secondary winding wrapped around the core; andwherein the core comprises at least one magnetic element operating in anon-saturated region (NSME).
 2. A transformer comprising: a core; aprimary winding wrapped around the core; at least one secondary windingwrapped around the core; and wherein the core comprises at least onemagnetic element operating in a non-saturated region (NSME).
 3. Thetransformer of claim 2, wherein said NSME further comprises a lowpermeability.
 4. The transformer of claim 3, wherein the NSME furthercomprises a mixture of 85% by weight of iron, 6% by weight of aluminum,and 9% by weight of silicon, thereby providing a wide thermal operatingrange for the magnetic element.
 5. The transformer of claim 3, whereinthe low permeability has a range of one to 500 u.
 6. The transformer ofclaim 5, wherein the NSME is an air magnetic element.
 7. The transformerof claim 2, wherein the NSME further comprises a B-H curvecharacteristic ranging from B=1 to 10,000 gauss and H=1 to 100 oersteds.8. A transformer comprising: a primary winding surrounding a pluralityof core elements; a secondary winding wrapped around a member of theplurality of core elements; and wherein at least one core elementcomprises a magnetic element operating in a non-saturated region (NSME).9. The transformer of claim 8, wherein said NSME further comprises a lowpermeability.
 10. The transformer of claim 9, wherein the NSME furthercomprises a mixture of 85% by weight of iron, 6% by weight of aluminum,and 9% by weight of silicon, thereby providing a wide thermal operatingrange for the magnetic element.
 11. The transformer of claim 9, whereinthe low permeability has a range of one to 500 u.
 12. The transformer ofclaim 11 wherein the NSME is an air magnetic element.
 13. Thetransformer of claim 8, wherein the NSME further comprises a B-H curvecharacteristic ranging from B=1 to 10,000 gauss and H=1 to 100 oersteds.14. A resonant sub-circuit comprising: a capacitor having a connectionto a magnetic element thereby forming a resonate relationship therebetween; and wherein said magnetic element further comprises a magneticelement operating in a non saturating region (NSME) thereof.
 15. Thesub-circuit of claim 14, further connected to a filter circuit, therebygiving the filter circuit a lower mass, complexity and cost for a givenfiltering characteristic.
 16. The converter of claim 15, wherein saidNSME further comprises a low permeability.
 17. The converter of claim16, wherein the NSME further comprises a mixture of 85% by weight ofiron, 6% by weight of aluminum, and 9% by weight of silicon, therebyproviding a wide thermal operating range for the magnetic element. 18.The converter of claim 16, wherein the low permeability has a range ofone to 500 u.
 19. The converter of claim 16, wherein the NSME is an airmagnetic element.
 20. The converter of claim 16, wherein the NSMEfurther comprises a B-H curve characteristic ranging from B=1 to 10,000gauss and H=1 to 100 oersteds.
 21. The sub-circuit of claim 14, whereinthe connection is a parallel connection.
 22. The sub-circuit of claim14, wherein the connection is a series connection.
 23. In a powerconversion circuit having an energized magnetic element, the improvementcomprising: said magnetic element operating in a non saturating region(NSME) thereof; said magnetic element having a natural frequency; andsaid energizing power approximately matched to said natural frequencythereby providing a higher power density, an improved thermal stability,and an increased conversion efficiency.
 24. The converter of claim 23,wherein said NSME further comprises a low permeability.
 25. Theconverter of claim 24, wherein the NSME further comprises a mixture of85% by weight of iron, 6% by weight of aluminum, and 9% by weight ofsilicon, thereby providing a wide thermal operating range for themagnetic element.
 26. The converter of claim 24, wherein the lowpermeability has a range of one to 500 u.
 27. The converter of claim 24,wherein the NSME is an air magnetic element.
 28. The converter of claim24, wherein the NSME further comprises a B-H curve characteristicranging from B=1 to 10,000 gauss and H=1 to 100 oersteds.
 29. Aninductor comprising: a core; a winding; and wherein the core has amagnetic element operating in a non-saturated region (NSME).
 30. Aninductor comprising: a plurality of cores; a winding around at least onemember N of the plurality of cores; and wherein the member N has amagnetic element operating in a non-saturated region (NSME).